Decorrelating Discrimination System of Code Division Multiple Access Signals

ABSTRACT

here is provided a new multi-user receiver configuration technique which solves the problems of the user signal separation function of the multi-user receiver such as the interference canceller, decorrelator, and least square error detection. That is, there is provided a code-division multiplex signal decorrelation/identification method for realizing much more excellent performance than the aforementioned methods in the aspect of the power band amplitude product required for one-bit transfer as the evaluation reference of the CDMA method. A function for successively judging the best user is added to the conventional least square error detector MMSE-D shown by a dotted line in the block diagram of the successive detection type least square error detection method (SD-MMSE).

TECHNICAL FIELD

The present invention relates to a code division multiple access communications system (CDMA) using spread-spectrum modulation which can reduce white noise admixed in a transmission process and interfering noise generated in a multi-user signal separation process by a receiver, can enhance the frequency-utilization-efficiency, and can reduce a power-bandwidth-product. In this case, the modulation/demodulation technology for a transceiver of a mobile communications system where the spread-spectrum modulation is applied to BPSK signals is taken as an example to explain user-separating techniques for a multi-user receiver.

BACKGROUND ART

Spread-spectrum communications is a system using spreading modulation technology where spreading-sequences are modulated by transmit-data. Due to this spreading modulation, a data-sequence spectrum having a relatively narrow bandwidth is spread to a wide frequency-band, producing a spread spectrum signal to be transmitted. In a region (cell or sector) where a base-station (BS) provides communications services, there are users of a plurality of user-stations (hereafter called users). Such a communications system is excellent in that a low transmission power per unit frequency is consumed, disturbance to other communications can be kept at a relatively low level, and the system has inherently strong resistance to jumming noise (AWGN) mixed in a transmission process and inter-station-interference-noise incoming from mobile stations other than a desired station, namely interfering stations.

However, since communications from a large number of stations share the same time-slot and the same frequency band, there is a problem in which an increase in the number of users to be accommodated per unit band is impeded by the inter-station-interference(-noise). That is to say, disturbance caused by such noise decreases frequency-utilization-efficiency and increases required transmit-power.

FIG. 16 is a block diagram illustrating the general construction of a mobile communications system which performs direct-sequence spread-spectrum (DS-SS) communications via a radio communications channel. Here, a transmitter TX_(k) of the k-th user u_(k) (k=1, 2, . . . K) among K users in a cell modulates a radio-band carrier-wave with binary transmit-data b_(k) to obtain a Binary Phase Shift Keying (BPSK) symbol s_(kBP), and modulates the k-th spreading-sequence g_(k) among K sequences allocated to K users with BPSK symbol s_(kBP) to produce a spread spectrum symbol s_(k). (symbol denotes a time limited signal conveying data) Thereafter, s_(k) is transmitted through a radio communications channel. In order to discriminate the addresses of the K users, pseudo-noise (PN) sequences each of which is different from one another are used as the k-th sequence g_(k).

A receiver RX receives through an antenna a receive-symbol r which includes, as the components, spread-spectrum-modulated symbol received from all the users, and demodulates receive-symbol r by a local carrier-wave {circumflex over (f)}₀ (=f₀) to obtain a base-band symbol r_(BB). Receiver RX applies the base-band symbol to a matched filter MF_(k) matched to the k-th spreading-sequence g_(k) to generate a soft-output {tilde over (b)}_(k) as the k-th soft-output. Soft-output {tilde over (b)}_(k) is compared with a threshold value by a hard decision circuit DEC to obtain the k-th detected value of binary data {circumflex over (b)}_(k) (the k-th means that the data has been sent from the k-th user) (This matched filter detection is called “correlative detection”).

Detected data {circumflex over (b)}_(k) is applied to a synchronizing circuit SYNC. A generating timing of the spreading-sequence is controlled so as to be synchronized with the carrier phase and the component of the k-th user specific received symbol component contained in receive-signal r. In TX and RX in FIG. 16, the arrangement of sequential order of multiplying functions of carrier-wave f_(C)({circumflex over (f)}_(C)) and spreading sequence g_(k) are often exchanged each other. However, the overall modulation and demodulation-functions remain the same, and any configuration may be used.

The above-described receiver uses a reception system where different respective matched filters to detect corresponding user symbols are arranged in parallel.

In this system, a cross-correlation between the k-th sequence g_(k) allocated to a user and the k′-th (different) sequence g_(k′)(k≠k) allocated to another user cannot be designed to be kept at a sufficiently low level, when the number K of users increase. A pilot-response p_(k) is influenced by a multipath channel gain between transceivers in addition to the spreading sequence, and an inter-user cross-correlation between a pair of such pilot responses takes a larger value than the correlation between the corresponding spreading-sequences themselves. Furthermore, the multipath delayed waves of an adjacent symbol generate an inter-symbol interference (ISI), which impedes an increase in the number K of users. Hance, it is impossible to improve the frequency-utilization-efficiency.

In order to suppress disturbance caused by the above-described interfering noise, many methods for multi-user receivers which perform user separation and adjacent symbol separation by solving decorrelating equations have been studied. However, a sufficient noise suppression effect has not been obtained. Here, a list of seven preceding techniques closely related to the present invention is shown below.

(A) [Mamoru Sawahashi, Yoshinori Miki, Hidehiro Andoh and Kenichi Higuchi, Pilot Symbol-Assisted Coherent Multistage Interface Canceller Using Recursive Channel Estimation for DS-CDMA Mobile Radio” IEICE Trans. Commun., Vol.E79-B, No. 9, pp. 1262 to 1270, (09.1996)]

(B) [Mitsuhiro Tomita, Noriyoshi Kuroyanagi, Satoru Ozawa and Naoki Suchiro, “Error rate performance improvement for a de-correlating CDMA receiver by introducing additional dummy pilot response”, PIMRC '02, Lisbon (09.2002)]

(C-1) [D. Koulakiotis and A. H. Aghvami, CTR, King's Collage, University of London “Data Detection Techniques for DS-CDMA Mobile Systems: A Review”, IEEE Personal Comm., pp. 24 to 34, June 2000]

(C-2) [T. Abe and T. Matsumoto, “Space-Time Turbo Equalization and Symbol Detection in Frequency Selective MIMO Channels with Unknown Interference”, Proc. WPMC '01, Aalborg Denmark (09.2001)]

(D) [Siavash M. Alamouti “A Simple Transmit Diversity Technique for Wireless Communications” IEEE JSAC, Vol. 16, No. 8 (10.1998)]

(E) [Naoki Seuhiro, Noriyoshi Kuroyanagi, Toshiaki Imoto and Shinya Matsufuji, “Very Efficient Frequency Usage Systems using Convolutional Spread Time Signals Based on Complete Complementary Code”, PIMRC '2000, (09.2000)]

(F) [Jiangzhou Wang and Jun Chen “Performance of Wideband CDMA Systems with Complex Spreading and Imperfect Channel Estimation” IEEE JSAC Vol. 19, No. 1, (01.2001)]

System (A) intends to upgrade the function of the k-th matched filter MF_(k) to detect a data of the k-th user u_(k) in the system explained with FIG. 16, and uses a receiver equipped with interference cancellers shown in FIG. 17. In an interference canceller IC-1 (the first stage), a matched filter bank MFB generates estimated transmit-data (soft-outputs) {tilde over (b)}_([k]) of all the users except that of the (k1)-th user by using a received input r¹ and a pilot-response supplied from a pilot response memory PRM. By using soft-outputs {tilde over (b)}_([k]), the first interference generator I-GEN₁ generates a replica (pseudo input) Φ_([k]). By subtracting Φ_([k]) from input r¹, interference canceller IC-1 generates a soft-output {tilde over (b)}_(k1). By making a hard decision on soft-output {tilde over (b)}_(k1), is obtained a detected value {circumflex over (b)}_(k), with which a corresponding replica Φ_(k1) is generated with the second interference generator I-GEN₂. To a canceller (called the second stage) IC-2, is applied an input r² which is made by subtracting replica Φ_(k1) from received input r¹. Canceller IC-2 repeats to apply the same operation to input r² as that IC-1 has done. In this method, due to existence of large cross-correlations between pilot-responses of respective users, large interference components resultantly remain in the soft-outputs. For this reason, an error rate cannot be sufficiently reduced.

FIG. 18 shows a functional block diagram of a multi-user receiver. FIG. 18( a) shows a De-correlating Detector (system DD) corresponding to System (B). In this case, each user transmitter transmits a pilot symbol, so as not to be disturbed by interfering waves from the other users. The receiver receives these pilot symbols and always prepares highly accurate pilot-responses from all the users in a pilot response memory PRM. Each user transmitter uses both a sequence allocated to the user, and a common carrier-wave used by all the users to generate a transmit-data-symbol to be transmitted by the transmitter. More specifically, in System (B), each user transmits a pilot-signal to a base-station (BS) so that BS can accurately recognize channel-gain-characteristics (channel) from each user to the base-station. Therefore, base-station BS can obtain a pilot-response (channel-gain-property) p_(k) of a transmission-paths from the k(=1, 2 . . . , K)-th user u_(k). A receive-symbol r is given by the following equation:

$\begin{matrix} {r = {{\sum\limits_{k = 0}^{K}\; {b_{k}P_{k}}} + x}} & \left( {A\text{-}1} \right) \end{matrix}$

where b_(k) is a transmit-data of the k-th user u_(k), and x is white noise (AWGN) included in receive-symbol r. By using a pilot-response-matrix P consisting of pilot-responses p_(k) of all the users, Eq. (A-1) is solved by a de-correlating detector (Decor) in FIG. 18 to obtain a soft-output {tilde over (b)}_(k)=b_(k)+Δb_(k) corresponding to the transmit-data where Δb_(k) is an error. This system has an advantage such that the influence of interfering waves can be completely removed.

However, in System (B), since pilot-response-matrix P is dependent on a channel gain, the regularity of matrix P often decreases, and the AWGN component is amplified in a process of solving the equation, resulting in an increase in an error Δb_(k) contained in the soft-output. Therefore, the number of users K and the spreading-sequence length L must satisfy a relationship: K<<L to reduce error Δb_(k). More specifically, there are problems such that the number of users to be accommodated is limited and the system is forced to have a low frequency-utilization-efficiency.

Systems (C-1) and (C-2) use a Minimum-Mean-Square-Error Detector (MMSE-D) shown in FIGS. 18( b) and 18(c). Although system MMSE-D provides a method to solve the same de-correlating equations as solved by system DD to generate soft-outputs, in order to increase the regularity of matrix P, it modifies matrix P with an additive term in the solving process. Due to the additive term, interfering noise is generated. Since the interfering noise decreases signal components contained in a receive-vector, the quality of the soft-output are degraded. System (C-2) provides a method to overcome a drawback of the MMSE-D. Consider the k-th user u_(k) as a target user. A soft-output-vectors {tilde over (b)} consisting of all the user components is calculated by a conventional (as the first stage) MMSE, and an estimated received input (replica)  _([k]) received from users except the k-th user u_(k) is calculated from a channel matrix H composed of channel gains of all the users estimated from the pilot-response P, and a soft-output-vectors {tilde over (b)}_([k]) which is made by removing the k-th user's soft-output component from soft-output-vector {tilde over (b)}, The receiver calculates the second stage receive-symbol r^(k) by removing replica φ_([k)] that is an interference component for the k-th user u_(k), from the first stage receive-symbol r¹. Symbol r^(k) is applied to a conventional (as the second stage) MMSE, to obtain the k-th soft-output {tilde over (b)}k. The system makes hard decision on soft-output {tilde over (b)}_(k) to produce the k-th detected value {circumflex over (b)}_(k). Since soft-output-vector {tilde over (b)}_([k]) does not include the k-th soft-output {tilde over (b)}_(k), an interference component generated by the k-th data b_(k) cannot be removed in this process. Therefore, since replica φ_([k]) includes large interfering noise, an improvement in frequency efficiency is not sufficient.

System (D) deals with a single-user system where space-time coded transmission is performed on a multiple-input multiple-output (MIMO) system. In the system, by using a plurality (N) of transmit-antennae as a space axis and a plurality (N_(τ)) of symbol slots as a time axis, N data are transmitted using NN_(τ) symbols to get an advanced space-time diversity effect. In order to design respective transmit-symbol sets each consisting of N symbols simultaneously to be transmitted so as to have orthogonalities each other, a transmitter multiplies the respective of N transmit-symbols of the respective combinations by respective element sequences (code-words) of an orthogonal sequence-set (code) so that a receiver easily can separate the N data from each other. An effective technique such as to apply this method to a multi-user receiver has not been established. Furthermore, in the system, since this system assumes that the channel characteristics during a time for transmitting N_(τ) symbols are invariant, a sufficient diversity effect on the time axis cannot be obtained.

System (E) utilizes such a characteristic that complementary sequences have orthogonalities at all shift positions. A user of this system produces a synthesized transmit-symbol by adding plural element sub-symbols, each is made so as to multiply one of the sequentially chip shifted complementary sequences by transmit-data and transmits the synthesized transmit-symbol. Since this system can transmit a large number of element sub-symbols over a symbol period, frequency-utilization-efficiency is improved. Furthermore, the receiver can easily separate and discriminate the plural data carried on sub-symbols because the shifted sequences are orthogonal each other. However, since a sum of a large number of the element sub-symbols is transmitted, there is a problem such that a peak transmit-power considerably increases.

System (F) is a pilot-data multiplexed transmission system standardized as one of the 3^(rd) Generation systems in which a user transmits simultaneously a pilot-symbol and a data-symbol. In order to separate a data and a pilot at a receiver, this system sets a real number as data (b ε±1) and an imaginary number a pilot (p=j).

The same spreading-sequence is modulated by a complex number (b+j) to make a transmit-symbol. More specifically, since an 1-bit data/symbol is transmitted by using a symbol time slot and a band-width in which a 2-bit/symbol can be transmitted by QPSK (4-phase shift keying modulation), the pilot consumes considerably large resources equal to that the data dose. Furthermore, since a pilot is subjected to interferences by both a data symbol from the same user and pilot/data-symbols from the other users due to the multi-path channel characteristics, there is also a problem such that an accurate pilot-response cannot be easily obtained.

DISCLOSURE OF THE INVENTION

The present invention has been made to overcome disadvantages which conventional user signal separating functions of multi-user receivers have. As the conventional techniques, the interference canceller, the de-correlating detector, and the minimum-mean-square-error detector have been examined. The object of this invention is to provide design techniques of new multi-user receivers, and thereby to realize considerably more enhanced performance than that of the above described conventional systems with respect to a power-bandwidth-product required for 1-bit transmission that is an evaluation measure of performance of CDMA systems.

In order to achieve the above object, according to an invention described in claim 1, there is provided a decorrelating discrimination system of code division multiple access signals, wherein a basic system structure is composed of plurality of cells, each of the plurality of cells comprises a base-station and K user-stations, each of the user-stations including a user transmitter and a user receiver, communicating through a multi-access-channel with the base-station which includes a base-station receiver and a base-station transmitter, and the user transmitter is capable of transmitting a data symbol to convey a data with a spreading sequence, and a pilot-symbol that is the spreading sequence to identify a channel from the user transmitter to the base-station receiver, and the base-station receiver includes a minimum mean square error detector to analyze an input vector, that is a receive-symbol containing both multiple user specific data responses, each having conveyed a transmit-data through a channel, in a way such that the minimum mean square error detector solves a system of linear equations with multiple unknowns made for the input vector, composed of a user separating matrix U consisting of a pilot matrix associated with the channels and a white noise power multiplied identity matrix, and an unknown data vector, characterized in that the receiver comprises: means for solving a system of equations as identified by the first system of decorrelating equations with a user separating matrix U⁰ to produce a soft-output vector {tilde over (b)}⁰ at an analyzing circuit, means for producing a variance of user-corresponding noise evaluation vectors of a noise evaluation matrix C generated by a matrix inverse of the user-separating matrix U as a correct solution measure P_(C) ⁰ consisting of K components at a power estimator, means for deciding one soft-output component {tilde over (b)}_(k′) ⁰ of the soft-output-vector {tilde over (b)}⁰ as the first best user u_(k), based on one of the minimum candidate components of the correct solution measure P_(C) ⁰ at a best user decision circuit, means for making a hard decision on the soft-output {tilde over (b)}_(k′) ⁰ to obtain a detected value {circumflex over (b)}_(k′) ⁰ at a decision circuit, means for removing components corresponding to the first best user u_(k′) from the first system of decorrelating equations with the user separating matrix U⁰ with circuits of a modulator, a subtractor, and a best user remover to generate a system of equations as identified by the second system of decorrelating equations with a user separating matrix U¹, means for solving the second system to produce a soft-output vector {tilde over (b)}¹, producing a variance of user-corresponding noise evaluation vectors of a noise evaluation matrix C generated by a matrix inverse of the user separating matrix U¹ as a correct solution measure P_(C) ¹ consisting of (K-1) components, and deciding one output {tilde over (b)}_(k′) ¹ of the soft-outputs of the soft-output-vector {tilde over (b)}¹ as the second best user u_(k″) based on one of the minimum candidate components of the correct solution measure P_(C) ¹, means for making a hard decision on the soft-output {tilde over (b)}_(k′) ¹ to obtain a detected value {circumflex over (b)}_(k′) ¹, means for sequentially repeating the same method as that applied to the second system of decorrelating equations to the following systems of decorrelating equations, to decide the following best users, thereby producing the best users in turn, and means for making hard decision on soft-outputs of the best users to obtain detected values of transmit-data all the users have sent out at the decision circuit.

According to an invention described in claim 2, there is provided a decorrelating discrimination system of code division multiple access signals, according to claim 1, characterized in that the receiver comprises: means for solving a system of equations as identified by the first system of decorrelating equations with a user separating matrix U⁰ to produce a soft-output vector {tilde over (b)}⁰ at an analyzing circuit, means for producing the variance of user-corresponding noise evaluation vectors P_(C) ⁰ consisting of K components at a power estimator, obtaining input noise power N_(r0) at a noise power estimator, and calculating a standard deviation σ⁰ to compose an error amplitude distribution with the variance P_(C) ⁰ and the input noise power N_(r0), means for obtaining a ratio identified by the 0-th normalized probability ratio λ⁰ consisting of K components that is calculated based on an error distribution model with the standard deviation σ⁰ and the K components of the soft-output vector {tilde over (b)}₀, means for deciding one soft-output component {tilde over (b)}_(k′) ⁰ of the soft-output-vector {tilde over (b)}⁰ as the first best user u_(k′) based on one of the maximum candidate components of normalized probability ratio λ⁰ at a best user decision circuit, means for making a hard decision on the soft-output {tilde over (b)}_(k′) ⁰ to obtain a detected value {circumflex over (b)}_(k′) ⁰ at a decision circuit, means for removing components corresponding to the first best user u_(k′) from the first system of decorrelating equations with the user separating matrix U⁰ according to claim 1, means for sequentially repeating the same method as that applied to the first system of decorrelating equations to the following systems of decorrelating equations, according to claim 1, and means for making hard decision on soft-outputs of the best users to obtain detected values of transmit-data all the users have sent out at the decision circuit.

According to an invention described in claim 3, there is provided a decorrelating discrimination system of code division multiple access signals, wherein a basic system structure is composed of plurality of cells, each of the plurality of cells comprises a base-station and K user-stations, each of the user-stations including a user transmitter and a user receiver, communicating through a multi-access-channel with the base-station in the cell which includes a base-station receiver and a base-station transmitter, and a user transmitter is capable of transmitting a data symbol to convey a data with a spreading sequence, and a pilot-symbol that is the spreading sequence to identify a channel from the user transmitter to the base-station receiver, and the base-station receiver includes a minimum mean square error detector to analyze an input vector, that is a receive-symbol containing both multiple user specific data responses, each having conveyed a transmit-data through a channel, in a way such that the minimum mean square error detector solves a system of linear equations with multiple unknowns made for the input vector, composed of a user separating matrix U consisting of a pilot matrix associated with the channels and a white noise power multiplied identity matrix, and an unknown data vector, characterized in that the receiver comprises: means for solving a system of equations as identified by the first system of decorrelating equations with a user separating matrix U⁰ identified by the 0-th user separating matrix to produce a soft-output vector {tilde over (b)}⁰ identified by the 0-th soft-output vector at an analyzing circuit, means for multiplying the 0-th soft-output-vector {tilde over (b)}⁰ by a matrix inverse of the 0-th user-separating matrix U⁰ to calculate an interference-correcting vector c⁰ identified by the 0-th interference-correcting vector at an interference generator, and adding the 0-th interference-correcting vector c⁰ to the 0-th soft-output-vector {tilde over (b)}⁰ to produce a soft-output-vector {tilde over (b)}¹ identified by the first soft-output-vector, means for applying the same method to calculate an interference-correcting vector c¹ identified by the first interference-correcting vector using the first soft-output-vector {tilde over (b)}¹ as that used for calculating 0-th interference-correcting vector c⁰, means for applying and makes hard decisions on respective components of a soft-output-vector {tilde over (b)}^(n) of the n-th stage calculated by repeating a method of adding the first interference-correcting vector c¹ the 0-th soft-output-vector {tilde over (b)}⁰ to produce a soft-output vector {tilde over (b)}² identified by the second soft-output vector, means for n times repeating the same method as that applied to obtain the second soft-output vector {tilde over (b)}² to produce the n-th soft-output {tilde over (b)}^(n), and means for making hard decision on soft-outputs of the n-th soft-output {tilde over (b)}^(n) to obtain detected values of transmit-data all the users have sent out at the decision circuit.

According to an invention described in claim 4, there is provided a decorrelating discrimination system of code division multiple access signals, according to claim 3, characterized in that the receiver comprises: means for introducing a coefficient λ_(N) to increase an amplitude of the identity matrix, used in the user-separating matrix U, and producing a system of decorrelating equations with a user-separating matrix U modified the coefficient λ_(N), means for limiting the amplitude of the soft-output-vector {tilde over (b)}⁰ of the 0-th stage calculated as a solution the system to produce a modified soft-output vector, generating an interference-correcting vector c⁰ of the 0-th stage by multiplying the modified soft-out vector by a matrix inverse of the user-separating matrix U, soft-out vector, means for adding a vector obtained by multiplying the interference-correcting output c⁰ by an interference power estimated coefficient θ to the soft-output-vector {tilde over (b)}⁰ of the 0-th stage to generate a soft-output {tilde over (b)}¹ of the first stage, means for repeating the same method to the following stages to obtain a soft-output-vector {tilde over (b)}^(n) of the n-th stage, and, means for making hard decision on soft-outputs of the n-th soft-output {tilde over (b)}^(n) to obtain detected values of transmit-data all the users have sent out at the decision circuit.

According to an invention described in claim 5, there is provided a decorrelating discrimination system of code division multiple access signals, according to claim 1 or 3, characterized in that the receiver comprises: means for receiving pilot-response-vectors received from respective user transmitters and separating each of them as a main response of a current pilot-symbol arrived on a target symbol-period and delayed wave responses of preceding pilot-symbols arrived on the same target symbol-period, and producing a pilot-response-set for each user, consisting of synthesized pilot-responses made by taking an algebraic sum of the main response and the delayed wave responses, means for generating a pilot-response-matrix P composed of the synthesized pilot-responses of all the users, generating a system of decorrelating equations with a user-separating matrix U made by the pilot-response-matrix P and an identity matrix, an unknown data-vector b, and receive-symbol-vector r as constituent elements, and solving the system according to a method of claim 1 or 3 to obtain a soft-output-vector.

According to an invention described in claim 6, there is provided a decorrelating discrimination system of code division multiple access signals, according to claim 1 or 3, characterized in that, the basic system comprising: means for including a multiple-input multiple-output system in which a plurality of antennae are arranged to perform communications, each of the user transmitters comprising: means for allocating a plurality (N_(d)) of transmit-data to N_(τ)N symbols on a space-time transmit-axis constituted by a plurality (N_(τ)) time slots and a plurality (N) of transmit-antennae, and transmitting N_(τ)N symbols over N_(τ) symbol periods, and the base-station receiver comprising: means for receiving symbols over N_(τ) symbol slots at a plurality (M) of antennae, storing a pilot-response P_(dτnm) ^(k) of a pilot-symbol received at the m-th receive-antenna when the k-th user transmitter sends d-th transmit-pilot-symbol of N_(d) symbols over the τ-th symbol-slot of N, symbol-slots, generating a concatenated pilot-response-vector P_(d) ^(k) made by concatenating only pilot-responses p_(dτnm) ^(K) corresponding to the d-th pilot-responses with respect to antenna number m and time-sequence numbers τ, generating a pilot-response-matrix P consisting of these vectors, and generating a concatenated receive-vector r made by concatenating of M pieces of receive-symbol-vectors received on the N_(τ) symbol slots, means for generating a system of decorrelating equations with a user-separating matrix U generated from the pilot-response-matrix P and an identity matrix, the concatenated receive-vector r, and an unknown-data-vector b, means for solving the system of decorrelating equations according to claim 1 or 3 to obtain a soft-output vector {tilde over (b)} of the transmit-data-vector b, and making {tilde over (b)} hard decisions on respective components of the soft-output-vector {tilde over (b)} to obtain a detected data vector {circumflex over (b)}.

According to an invention described in claim 7, there is provided a decorrelating discrimination system of code division multiple access signals, according to claim 6, characterized in that each of the user transmitters comprises: means for interleaving in advance a time sequence of N transmit-symbols where N is the number of transmit-antennae, and transmiting interleaved symbols over N_(τ) times, and the receiver comprises: means for performing deinterleaving M pieces of receive-symbols where M is equal to the number of receive-antennae, means for generating a system of decorrelating equations for each of N_(τ) symbol sets made by deinterleaved outputs, solving the system to obtain a soft-output-vector {tilde over (b)} of a transmit-data-vector b according to claim 6, and making hard decisions on respective elements of the soft-output-vector {tilde over (b)} to obtain a detected data-vector {circumflex over (b)}.

According to an invention described in claim 8, there is provided a decorrelating discrimination system of code division multiple access signals according to claim 1 or 3, wherein a basic system structure is composed of plurality of cells, each of the plurality of cells comprises a base-station and K user-stations, each of the user-stations including a user transmitter and a user receiver, communicating through a multi-access-channel with the base-station which includes a base-station receiver and a base-station transmitter, and characterized in that the user transmitter comprises: means for transmitting a data symbol to convey a data with a spreading sequence, and a pilot-symbol that is the spreading sequence to identify a channel from the user transmitter to the base-station receiver, means for generating an enveloped cyclically shifted spreading-sequence made by adding guard sequences to a core-spreading-sequence which belongs to a k-shift sequence of one pair of complete complementary spreading-sequences or a k-shift sequence of a zero correlation zone spreading-sequence as the core-spreading-sequence, means for controlling the transmit-timing so that all of user specific receive-symbol components may arrive at the base-station receiver under a synchronous or quasi-synchronous condition, and the receiver comprises: means for extracting a core-period-part of the receive-symbol as an input vector, and analyzing it with a minimum mean square detector according to a method of claim 1 or 3.

According to an invention described in claim 9, there is provided a decorrelating discrimination system of code division multiple access signals, according in any one of claims 1 to 8, characterized in that a user transmitter identified by the k-th user transmitter of K user transmitters comprises; means for generating a pilot-symbol with a guard added spreading-sequence, and preparing a pilot-symbol-sequence consisting of N symbols modulated by the k-th code-word with a code length N in an orthogonal code and transmitting the pilot-symbol-sequence so that it may arrive at the receiver together with other pilot-symbol-sequences sent out by the other user-stations under a synchronous or quasi-synchronous condition, and the base-station receiver comprises: means for receiving a pilot-response-sequence multiplexed by all of user specific pilot-responses, and applying the pilot-response-sequence to a matched filter matched to the k-th orthogonal code-word to generate a pilot-response-vector of the k-th user, and, means for producing a pilot-response-matrix P composed of pilot-response-vectors of all the K users to establish a the system of decorrelating equations used for respective claims 1 to 8.

According to an invention described in claim 10, there is provided a decorrelating discrimination system of code division multiple access signals, according to any one of claims 1 to 4, characterized in that the receiver comprises: means for solving a system of equations as identified by the first system of decorrelating equations with a decorrelating detector which is made by removing an identity matrix I from a user separating matrix U used in the minimum mean square error detector and, means for solving the following systems of decorrelating equations with decorrelating detectors.

According to an invention described in claim 11, there is provided a decorrelating discrimination system of code division multiple access signals, according claim 1 or 3, wherein the basic system comprising: means for including a multiple-input multiple-output system in which a plurality of antennae are arranged to perform communications, each of the user transmitters comprising: means for transmitting a data symbol to convey a data with a spreading sequence, and a pilot-symbol that is the spreading sequence to identify a channel from the user transmitter to the base-station receiver, and the base-station receiver comprising: means for receiving symbols over N_(τ) symbol slots at a plurality (M) of antennae, and characterized in that the receiver comprising: means for receiving M pieces of pilot-response-vectors per user obtained through M pieces of the antennae, generating an extended pilot-response-vector by concatenating the pilot-response-vectors and generating a pilot-response-matrix P by composing extended pilot-response-vectors obtained for all the users, means for generating an extended receive-vector r by concatenating all of the receive-symbols through M pieces of the antennae, stablishing a system of decorrelating equations with a user separating matrix U made by the pilot-response-matrix P and solving the system to obtain a soft-output vector according to claim 1 or 3.

According to an invention described in claim 12, there is provided a decorrelating discrimination system of code division multiple access signals, according to claim 5, characterized in that each of the user transmitters comprises: means for generating a data and a pilot-symbols with an extended sequence which is produced by adding an imitated delayed sequence to a core-spreading-sequence, so that the imitated delayed sequence is arranged outside the tail of a transmit-symbol-period that is the same time-slot as the core-spreading-sequence, and transmitting the data and the pilot-symbols so that a component corresponding to the imitated delayed sequence takes a time position overlapping a front portion of a subsequent symbol, transmitting a data-symbol and a pilot-symbol, and the receiver comprises: means for obtaining a receive-data-symbol and K user pilot-responses, and establishing a system of decorrelating equations with a user separating matrix or a pilot-response-matrix having an enhanced regularity, produced based on the receive data symbol and K user pilot-responses, and means for solving the system according to claim 5.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a transceiver.

FIG. 2 is a diagram showing a transmission-paths model.

FIGS. 3( a) and 3(b) are time charts of symbol-sequences composed of base-band transmit-/receive guard added symbols.

FIG. 4 is an illustration showing frequency distribution characteristics to calculate a correct solution occurrence probability based on a soft-output.

FIG. 5 is a block diagram of a sequential detection type MMSE receiver.

FIG. 6 is a block diagram of an interference-control MMSE detector.

FIG. 7 is a time chart showing base-band transmit-/receive-symbol-sequences and pilot-responses.

FIGS. 8( a) and 8(b) are diagram showing a method of generating positive/negative pilot-response-vectors.

FIG. 9 is a diagram showing transmit-symbols and pilot-responses of an imitated delayed wave transmission system.

FIG. 10 is a block diagram of a multi-user transmitter/receiver using a space-time coding system.

FIG. 11 are diagrams showing a signal generating model of the space-time coding system, in which FIG. 11( a) is an arrangement diagram of space-time symbols and FIG. 11( b) is a write-read order diagram for interleaving.

FIG. 12 is a time chart of transmit-/receive-symbols for a CDMA transmission system using complementary sequences.

FIG. 13 is a block diagram of a receiver of a CDMA transmission system using complementary sequences.

FIG. 14 is a time chart of transmit-/receive-symbol-sequences of an orthogonal sequence modulated pilot transmission system.

FIG. 15 is a block diagram of a transceiver of the orthogonal sequence modulated pilot transmission system.

FIG. 16 is a functional block diagram of a transceiver for a conventional CDMA communications system.

FIG. 17 is a function block diagram of a multi-user receiver (interference canceling system).

FIG. 18 are functional block diagrams of multi-user receivers (using decorrelating equations), in which FIG. 18( a) is a diagram showing a derrelator (DD), FIG. 18( b) is a diagram showing a minimum mean square error detector (MMSE-D), and FIG. 18( c) is a diagram showing a soft-output canceling minimum mean square error detector (SC-MMSE-D).

BEST MODE FOR CARRYING OUT THE INVENTION

FIG. 1 is an auxiliary explanatory diagram of the first embodiment of the present invention, and is a block diagram showing a configuration of a transceiver of a code division multiple access (CDMA) communications system. In FIG. 1, the k-th (k=0, 1, 2, . . . K) transmitter TX_(k) used by the k-th user u_(k) is shown on the left, and a receiver RX used in a base-station is shown on the right. A binary data b_(k)(ε±1) with symbol-period T_(E) is applied as an input to transmitter TX_(k). A symbol p denotes a pilot input which is set by p=1. Pilot input p is transmitted in a time-division-manner at a timing so as not to temporally overlap both transmit-data of all users and pilots transmitted by the other users with a switch S₁ shown in FIG.(a). Input b_(k) modulates an enveloped sequence e_(k)(i) with guard (to be described later) at a modulator MOD to generate an impulse-sequence b_(k)e_(k)(i) with L_(E) chips. A convolutional multiplier CONV, convolutionaly modulates a chip-waveform q generated on a carrier-wave having a frequency f_(c) with respective impulses of the impulse-sequence described above. (In order to transmit each chip impulse on a bandwidth which is approximately equal to a chip rate f_(ch) without inter-chip interference, a spreading range T_(q) of chip-waveform q is set to T_(q)>>T_(c) with respect to a chip spacing T_(C).) Convolutional multiplier CONV outputs a transmit-symbol s_(k)(t) of the k-th user represented as (hereafter a time limited signal conveying data is expressed as a symbol):

$\begin{matrix} {{s_{k}(t)} = {b_{k}\left\{ {\sum\limits_{i = 0}^{L - 1}\; {{e_{k}(i)}{q\left( {t - {iT}_{C}} \right)}}} \right\} \sqrt{2}\cos \; 2\; \pi \; f_{C}t}} & (1) \end{matrix}$

where e_(k)(i) is an enveloped sequence with chip variable i, and q(t) is a chip-waveform.

The above equation expresses one symbol. (When an adjacent symbol is present, it is modified to a linear sum of equation obtained by shifting the above equation by integer times a symbol period T_(E).)

In another time region different from the data transmission time, a symbol s_(kP)(t) generated by setting b_(k)=p=1 in Eq. (1) is transmitted as a pilot(symbol). All the users transmit transmit-data-symbols at such timings that corresponding user specific receive-symbols almost simultaneously arrive at the receiver. This is a quasi-synchronous condition. An inter-user deviation among receive-timings is controlled by a base-station so as to be less than a guard period.

Let us explain a receiver composition. A received symbol on a data transmission period at the receiver is given by the following equation as a sum of transmit-signals of all the users:

$\begin{matrix} {{r(t)} = {{\sum\limits_{k = 1}^{k = K}\; {{s_{k}(t)}*{h_{k}(t)}}} + {x_{c}(t)}}} & (2) \end{matrix}$

where h_(k)(t) is a channel (gain) vector between the k-th user and a base-station, x_(c)(t) is an AWGN, and * is convolutional multiplication. Received symbol r(t) is converted into a base-band symbol r_(b)(t) by both a modulator MOD 2 with a local carrier-wave {circumflex over (f)}_(c) and a band-pass-filter BPF. (In MOD 2, complex components are actually demodulated by using orthogonal carrier-waves.) The continuous time waveform of base-band symbol r_(b)(t) is applied to a correlator which matches to the same waveform q(t) as the chip-waveform used in the transmitter, to produce chip impulse (discrete) received symbol r_(b)(i) with period T_(E). Received symbol r_(b)(i) is applied to a gate AND which extracts a central portion (with period T) of the received symbol with period T_(E) by the designation of a core-symbol extracting signal CEX, to generate a core receive-symbol r_(c)(i) with period T.

During a pilot transmission period, a switch S₂ is turned upward, so that core received-symbol r_(c)(i) is resultantly applied to a pilot-response memory PRM and stored therein. In general, the whole pilot transmission period is divided by K, so that each of the divided period is used for pilot transmission of the individual user. Thus, the k-th core received symbol is given by the following equation:

r _(c)(i)=p _(k)(i)=(p _(k1) , p _(k2) . . . , p _(kL))^(T)  (3)

and serves as a pilot-response-vector p_(k) of the k-th user consisting of L chips. (For convenience, a vector, and a matrix consisting of vectors may be expressed by bold-italic and bold notations, respectively.) An uppercase T is a transposing operator, and chip component p_(ki) takes a complex amplitude. Although vector p_(k) includes white noise, by receiving pilot-responses a plurality of times and calculating an average vector with them, the noise component can be neglected. A pilot-response-matrix with a size (L×K) for K users is given by the following equation:

P=[p₁ p₂ . . . p_(K)]  (4)

On the other hand, respective pilot-responses p_(k)(i) are applied to a noise power estimating circuit E(N_(r0)). Circuit E(N_(r0)) produces a sum of a large number of pairs of vectors p_(k)(i) while inverting the polarity of p_(k)(i) alternately, to obtain a mean squared value as noise power. On the basis of an average value, the same mean values for all the pilot-responses as done for the k-th pilot responses, an estimated value of noise power N_(r0) included in core-receive-symbol r_(c)(i) is obtained.

During a data transmission period, switch S₂ is connected downward, core-receive-symbol r_(c)(i) including white nose with power N_(R0) is resultantly applied to a detector MMSE-D to which a pilot-response-matrix P is supplied from pilot-response-memory PRM. Detector MMSE-D outputs a decision vector {circumflex over (b)}=({circumflex over (b)}₁, {circumflex over (b)}₂, . . . , {circumflex over (b)}_(k), . . . , {circumflex over (b)}_(K))^(T) corresponding to a transmit-data-vector b=(b₁, b₂, . . . , b_(k), . . . , b_(k))^(T).

FIG. 2 is an auxiliary explanatory diagram of the present invention, showing a model diagram of multiple access transmission channels. Outputs of the K users are multiplexed in the transmission process and arrive at a receiver. The k-th transmit-data-symbol s_(k)(t) and the k-th pilot-symbol s_(kP)(t) the k-th user u_(k) has sent out are subjected to a conversion by a channel vector h_(k) from the transmitter to the base-station receiver. These symbols generate a base-band receive-data-symbol r_(k)(i) and a pilot-symbol r_(kP)(i) as given by the following equations:

$\begin{matrix} \left. \begin{matrix} {{r_{k}(i)} = {{{s_{k}(i)}*{h_{k}(i)}} = {\sum\limits_{j = 0}^{J - 1}\; {{s_{k}\left( {i - j} \right)}h_{kj}}}}} \\ {{r_{kP}(i)} = {{{s_{kP}(i)}*{h_{k}(i)}} = {\sum\limits_{j = 0}^{J - 1}\; {{s_{kp}\left( {i - j} \right)}h_{kj}}}}} \end{matrix} \right\} & (5) \end{matrix}$

where h_(k)=(h_(k0), h_(k1), . . . , h_(kj), . . . , h_(kJ-1))^(T) is a complex channel vector consisting of J chips with chip-period T_(C) which is equivalent to a channel resolution. Note that J is the number of multi-path waves including a direct wave and delayed waves.

FIG. 3 is an auxiliary explanatory diagram of the first embodiment of the present invention, showing a timing relationship between transmit-/receive-symbol sequences. When the k-th user u_(k) (k=1,2) transmits a transmit-symbol sequence as shown in FIG. 3( a), receive-symbol sequences received from the respective users are shown in user multi-path component wise in FIG. 3( b), where a user specific receive-symbol corresponding to individual transmit-symbol is composed of three multi-path components associated with a case of J=3. Consider a sum of the maximum value τ_(m) of a deviation between direct waves of a pair of user specific receive-symbols (τ_(m)=τ₁₂ for the case shown in FIG. 3) and a delayed wave spread (J−1)T_(c). According to the following expression:

T _(h) ,T _(l)>|τ_(m)|+(J−1)T _(C)  (6)

A guard sequence period (guard time) T_(h)(=L_(h)T_(C)) and T_(l)(=L_(l)T_(C)) should be designed so as to be longer than the sum. This condition is satisfied by controlling transmit-timing of all the users. This control provides a condition such that respective symbol boundaries F_(B) shown in FIG. 3 are not included in the core period T, that is a central portion of enveloped symbol-period T_(E), as shown by a time duration enclosed by double dotted lines. Thus let us extract a core-receive-symbol r_(c)(t) over core-period as an object to be analyzed, because due to the protecting function of the guard sequence (that is to avoid ISI), a simple analysis can be performed.

Here, a base-band received discrete symbol (over period T_(E)) associated with Eq. (2) is represented by the following equation:

$\begin{matrix} \left. \begin{matrix} {{r_{b}(i)} = {{\sum\limits_{k = 1}^{K}\; \left\{ {{s_{k}(i)}*{h_{k}(i)}} \right\}} + {x(i)}}} \\ \left\{ {{- L_{h}} \leq i \leq \left( {L + L_{i} - 1} \right)} \right\} \end{matrix} \right\} & (7) \end{matrix}$

where L_(h) and L_(l) are a header length and a tail length of the guard sequences, respectively. A core-symbol is given by limiting the chip-sequence-range of a symbol:

r _(C)(i)=r _(b)(i) {0≦i≦(L−1)}  (8)

When the enveloped spreading-sequence e_(k)(i) is defined as a periodic sequence of core-sequence g_(k)(i), the j-th delayed wave component of the core-sequence included in the core receive-symbol can be expressed as h_(kj)g*_(k)(i−j), where g*_(k)(i−j), is the j-shift sequence of g_(k)(i) and upper script * means a sequence with L chips. For this reason, r_(c)(i) is given by the following equation:

$\begin{matrix} {{r_{C}(i)} = {{\sum\limits_{k = 1}^{K}\; {\sum\limits_{j = 0}^{J - 1}\; {b_{k}h_{kj}{g_{k}^{*}\left( {i - j} \right)}}}} + {x(i)}}} & (9) \end{matrix}$

Symbol r_(kp)(i) in Eq. (5) is equivalent to data-symbol r_(C)(i) obtained by setting b_(k)=1 in Eq. (9). According to Eq. (3), the following equation holds good:

$\begin{matrix} {{r_{C}(i)} = {{{r_{S}(i)} + {x(i)}} = {{\sum\limits_{k = 1}^{K}\; {b_{k}{p_{k}(i)}}} + {x(i)}}}} & (10) \end{matrix}$

Therefore, the core receive-symbol is composed of a sum of binary value modulated user specific pilot-responses, and is given by any one of the following vector representations:

r _(c) =Pb+x  (11-A)

P{tilde over (b)}=r _(c) =r _(s) +x  (11-B)

where r_(c) is a core receive-(data)vector which includes a receive-signal vector r_(s) [first term of a right-hand side in Eq. (10)] and a white noise vector x=(x₁, x₂, . . . , X_(L)) as the components, b denotes a transmit-data-vector, {tilde over (b)} is a soft-output vector obtained by adding an error vector Δb to b, and P denotes a matrix in Eq. (4). When the both the side terms of Eq. (11-B) are multiplied by a Hermitian (transposed conjugate) matrix p^(H) of a pilot-response-matrix P,

P^(H)p{tilde over (b)}=P^(H)r_(C)  (12)

is obtained. This is a system of linear equations with multiple unknowns obtained when a decorrelating detector (DD) is used. In order to increase the regularity of P^(H)P of the above equation, a system of linear equations of the minimum mean square error (MMSE) criterion is used. The system is represented by replacing matrix P^(H)P in Eq. (12) with the following matrix U, according to a known theory. (Since this system is later used for a decision of the best user at the first stage, the system is called the first system of decorrelating equations.)

$\begin{matrix} \left. \begin{matrix} {{Ub} = {P^{H}{r_{C}\left( {= y} \right)}}} \\ {\left( {K \times K} \right)\left( {K \times 1} \right)\mspace{14mu} \left( {K \times L} \right)\left( {L \times 1} \right)} \\ {\left( {K \times K} \right)\left( {K \times 1} \right)\mspace{14mu} \left( {K \times L} \right)\left( {L \times 1} \right)} \end{matrix} \right\} & (13) \\ {\left. \begin{matrix} {U = {{P^{H}P} + {N_{r\; 0}I}}} \\ \begin{matrix} {\left( {K \times K} \right)\mspace{14mu} \left( {K \times L} \right)\left( {L \times K} \right)\mspace{14mu} \left( {K \times K} \right)} \\ \begin{pmatrix} \; & {\rho_{11} + N_{r\; 0}} & \rho_{12} & \ldots & \rho_{1\; K} \\ \; & \rho_{21} & {\rho_{22} + N_{r\; 0}} & \ldots & \rho_{2\; K} \\  = & \vdots & \vdots & \ddots & \vdots \\ \; & \rho_{K\; 1} & \rho_{K\; 2} & \ldots & {\rho_{KK} + N_{r\; 0}} \end{pmatrix} \end{matrix} \end{matrix} \right\} {\rho_{lk} = {p_{}p_{k}}}} & (14) \end{matrix}$

where U is a user-separating matrix having row-number l(=1, 2, . . . , L) and column-number k(=1, 2, . . . , K), I is an identity matrix, N_(r0) is power of white noise vector x, and y is a matched filter output vector (will be described later). The sizes (the number of rows×the number of columns) of a matrix and a vector are described in parentheses. Therefore, the system of equations are solved to produce a soft-output-vector {tilde over (b)} that is given by the following equations:

{tilde over (b)}=b+Δb=U ⁻¹ P ^(H) r _(C)  (15)

Δb=Δb _(x) +Δb ₁  (16-A)

Δb _(x) ≅U ⁻¹ P ^(H) x=Cx  (16-B)

Δb _(I) =−U ⁻¹ b  (16-C)

where Δb is an error vector consisting of a component Δb_(x)=[Δb_(x1), Δb_(x2), . . . , Δb_(xk), . . . . Δb_(xK)]^(T) depending on white noise x included in a received symbol, and an interfering noise component Δb_(l)=[Δb_(l1), Δb_(l2), . . . , Δb_(lk), . . . , Δb_(lK)]^(T) generated due to the additive term N_(r0)I, and C denotes a noise evaluation matrix (will be described later). Note that Eq. (16-B) includes a certain interfering noise component.

The k-th component {tilde over (b)}_(k) of the soft-output-vector obtained in Eq. (15) is made by a hard decision to obtain the k-th detected value {circumflex over (b)}_(k). Setting N_(r0)=0 in Eqs. (13) to (16) makes a relation U=P^(H)P, resulting in a system of equations for a system DD. In this case, interfering noise of Eq. (16-B) is eliminated, leading to Δb₁=0. However, Δb_(x) increases in comparison with a system MMSE. The above function is almost achieved by a conventional technology. However, since the conventional technology does not use guard sequences, it degrades the performance due to inter-symbol interference caused by delayed waves. Therefore, it is difficult to sufficiently decrease a bit error rate (BER) with the conventional methods. As the number of users K increases, the regularities of the matrices P^(H)P and U are deteriorated, and rate BER increases. As a consequence, the above described conventional systems are forced to have disadvantages such as an increase in transmit-power and reduction of frequency-utilization-efficiency. In order to solve the problem, the present invention uses the following method.

A sequential detection type CDMA multi-user reception system according to the first embodiment of the present invention is described below.

A right-hand-side matrix C in Eq. (16-B) is expressed by the following equation:

$\begin{matrix} \left. \begin{matrix} {C = {{U^{- 1}P^{H}} = \left\lbrack {C_{1},C_{2},\ldots \mspace{14mu},C_{k},\ldots \mspace{14mu},C_{K}} \right.}} \\ {C_{k} = \left( {c_{k\; 1},c_{k\; 2},\ldots \mspace{14mu},c_{kL}} \right)^{T}} \\ {c_{k\; l} = {\frac{\left( {k + l} \right)^{- 1}{p_{kl}^{*}\left\lbrack {\overset{\sim}{U}}_{kl} \right\rbrack}}{\det \lbrack U\rbrack}\left( {{l = 1},2,\ldots \mspace{14mu},L} \right)}} \end{matrix} \right\} & (17) \end{matrix}$

where C_(k) is called a noise evaluation vector corresponding to the k-th user, det [U] denotes the determinant of matrix U, and [Ũ_(kl)] denotes a cofactor obtained by removing the k-th row and the l-th column from matrix U. Here, using Eq. (16-B), let us obtain an evaluated average power P_(xk) of the k-th component Δb_(xk) of an error vector Δb_(x) included in the soft-output of the k-th user, by the following equations:

$\begin{matrix} {P_{xk} = {{E{{\Delta \; b_{xk}}}^{2}} = {{E\left\{ {\sum\limits_{l = 1}^{L}{\; {c_{kl}{\overset{\_}{x}}_{l}}}^{2}} \right\}} = {P_{Ck}{N_{r\; 0}/L}}}}} & (18) \\ {P_{Ck} = {\sum\limits_{l = 1}^{L}{\; c_{kl}}^{2}}} & (19) \end{matrix}$

Therefore, an expectation of evaluated average power P_(xk) is in proportion to a Power P_(Ck) of vector C_(k) that is the k-th component of matrix C. Thus, a power vector P_(C)=(P_(C1), P_(C2), . . . P_(CK)) is obtained. For this reason, as power P_(Ck) decreases, a correct solution probability of a hard decision value on the k-th soft-output {tilde over (b)}_(k) calculated for user u_(k) increases. [Since P^(H)x in Eq. (16-B) is considered as a noise input, even though P_(xk) is calculated by setting C=U⁻¹, almost the same evaluation can be obtained.]

FIG. 4 is an auxiliary explanatory illustration of the present invention, showing model characteristics of a soft-output distribution.

These characteristics are obtained by considering that the correct solutions are given as b_(k)ε±1 in Eq. (11) and an error voltage Δb_(xk) can be assumed to have a Gaussian distribution with the following standard deviation σ_(k):

$\begin{matrix} {\sigma_{k} = {\sqrt{\alpha_{k}N_{0}} = {\sqrt{\alpha_{k}{N_{r\; 0}/L}} = \sqrt{P_{xk}}}}} & (20) \\ {\alpha_{k} = {L{\frac{P_{xk}}{N_{r\; 0}}.}}} & (21) \end{matrix}$

where α_(k) is called a noise amplification factor, and N₀ is power of a data-symbol (not spread yet). From the actually obtained value of soft-output vector {tilde over (b)}, Log Likelihood Ratio LLR is obtained by the following equation with τ_(k) in Eq. (20). The correct solution probability is given by normalizing LLR as a normalized probability ratio λ_(k).

$\begin{matrix} \left. \begin{matrix} {{{LLR}\left( b_{k} \right)} = {{\log_{e}\frac{p_{r\; 1}\left( {b_{k} = {1{\overset{\sim}{b}}_{k}}} \right)}{p_{r\; 0}\left( {b_{k} = {{- 1}{\overset{\sim}{b}}_{k}}} \right)}} = \frac{p_{r\; 1}\left( \sigma_{k} \right)}{p_{r\; 0}\left( \sigma_{k} \right)}}} \\ {\lambda_{k} = {\tan^{h}{{LLR}\left( b_{k} \right)}}} \end{matrix} \right\} & (22) \end{matrix}$

where p_(r1)(σ_(k)) and p_(r0)(σ_(k)) are probability values obtained by assuming a Gaussian distribution with standard deviation assumes σ_(k) as shown in FIG. 4.

Normalized probability ratio λ_(k) has a value ranging from 1 to −1 and the polarity which is equal to that of the corresponding soft-output. As |λ_(k)| is close to 1, the correct solution probability of a hard decision value on the k-th soft-output {tilde over (b)}_(k) increases.

An arbitrary function γ_(k) including P_(ck) in Eq. (19) and γ_(k) in Eq. (22) as the elements can be used as a correct solution measure.

γ_(k) =F(P _(Ck),γ_(k))  (23)

When P_(Ck) or λ_(k) is directly used as a correct solution measure, the best user u_(k′) (k′ε 1, 2, . . . K) satisfies the following equation.

k′=arg min [P _(Ck)](kε1, 2, . . . , K)  (24-A)

k′=arg max [λ_(k)](kε1, 2, . . . , K)  (24-B)

As power P_(Ck′) decreases (as λ_(k′) increases), the error power of a soft-output {tilde over (b)}_(k′) of a user u_(k′) decreases and a correct solution probability of a detected value {circumflex over (b)}_(k′) of user u_(k′) increases.

In the above description, the k′-th user having the highest correct solution measure can be determined as the first best user k′. A method of determining the second best user will be described below. First, by removing the first best user component from both core-receive-symbol r, and pilot-response-matrix P, the following equation is obtained:

$\begin{matrix} \left. \begin{matrix} {r_{C}^{1} = {r_{C} - {{\hat{b}}_{k^{\prime}}p_{k^{\prime}}}}} \\ {P^{1} = \left( {p_{1}^{1},p_{2}^{1},\ldots \mspace{14mu},p_{k^{\prime} - 1}^{1},p_{k^{\prime} + 1}^{1},{\ldots \mspace{14mu} p_{K}^{1}}} \right)} \end{matrix} \right\} & (25) \end{matrix}$

When the above-equation is substituted into r_(c) and P in Eqs. (13) and (14), the following second system of decorrelating equations is obtained:

$\begin{matrix} \left. \begin{matrix} {{U^{1}b^{1}} = {P^{1\; H}r_{C}^{1}}} \\ {{{\left\lbrack {\left( {K - 1} \right) \times \left( {K - 1} \right)} \right\rbrack \left\lbrack {\left( {K - 1} \right) \times 1} \right\rbrack}\left\lbrack {\left( {K - 1} \right) \times L} \right\rbrack}\left( {L \times 1} \right)} \end{matrix} \right\} & (26) \\ \left. \begin{matrix} {U^{1} = {P^{1\; H}P^{1}}} \\ \begin{matrix} \begin{bmatrix} {\left( {K - 1} \right) \times} \\ \left( {K - 1} \right) \end{bmatrix} & {\left\lbrack {\left( {K - 1} \right) \times L} \right\rbrack \left\lbrack {L \times \left( {K - 1} \right)} \right\rbrack} & \begin{bmatrix} {\left( {K - 1} \right) \times} \\ \left( {K - 1} \right) \end{bmatrix} \end{matrix} \end{matrix} \right\} & (27) \end{matrix}$

Where b¹ is a transmitted data vector with all the user's components except one that is b_(k) ¹.

According to these equations, although the length of core-receive-symbol r¹ _(c) is invariant (length L), the size of matrix decreases from U⁰(K×K) to U¹{(K−1)×(K−1)} [U⁰ is of use for the first system in Eq.(14)]. In general, the decrease in size advantageously makes the regularity of matrix U¹ higher than that of U. Furthermore, since P¹ does not include p_(k′), a component ρ_(k′k)+N_(r0) included in U in Eq. (14) is eliminated, and a part of an interference vector Δb¹ generated by additive term N_(r0)I is eliminated. The improvement in regularity of matrix U decreases the absolute amplitude of error vector |Δb_(x)| in Eq. (16-B), and the removal of τ_(k′k)+N_(r0) included in matrix U decreases the absolute amplitude of error vector |Δb_(I)| in Eq. (16-C). For this reason, absolute error amplitude of error vector |Δb| generally decreases. When Eq. (26) is solved, the following equation is obtained:

{tilde over (b)} ¹ =b ¹ +Δb ¹ =[U ¹]⁻¹ p ^(1H) r _(C) ¹  (28)

Let a soft-output error vector in Eq. (16-A) be Δb⁰. Consider a comparison between soft-output error vectors Δb⁰ and Δb¹ in Eq. (28) with respect to power. For the above reason, the following equation is established:

|Δb ^(1|) ² <|Δd ⁰|²  (29)

Therefore, when detected value {circumflex over (b)}_(k′) of the first best user is correct, an average error-rate of (the detected data of) the users except for the first best user is lower than that of (that of) all the users calculated from Eq. (15). Equation (28) with which an improved error characteristic can be expected is solved, to determine the second best user k″ among the remaining users by the same method as that performed with Eq. (23) or (24). In this case, P_(Ck) in Eq. (19) must be again calculated by putting C¹=[U¹]⁻¹P^(1H) in place of C in Eq. (17). In this manner; respective errors included in both soft-outputs {tilde over (b)}_(k′) and {tilde over (b)}_(k′) obtained at the 0-th stage soft-output and the first stage are considerably smaller than an average value of errors of soft-outputs obtained from Eq. (15) in a conventional system. These processes are repeated, transmit-data of all the users are thus sequentially detected.

FIG. 5 is a block diagram of a sequential-detection-type minimum-mean-square-error detection system (SD-MMSE) showing the first embodiment of the present invention. In FIG. 5, a function of sequentially determining of the best users as described above is added to a conventional minimum mean square error detector MMSE-D indicated by a frame with dotted-line. An operation of detector MMSE-D will be described below. A core-receive-input r_(c) is applied to K matched filters MF(p_(k)) matched to pilot-responses p_(k) to generate correlation outputs γ_(k)(k=1, 2, . . . K), respectively.

A correlation vector y consisting of K pieces of the components γ_(k) is equivalent to the right-hand-side term of Eq. (13) and is given by:

y=(y ₁ , y ₂ , . . . , y _(k) , . . . , y _(K))^(T) =P ^(H) r _(C) +x  (30)

U-GEN in FIG. 5 denotes a generating circuit to generate matrix U expressed in Eq. (14), using both pilot-response-matrix P which is made with all the pilot-responses p_(k)(i) stored in pilot-response-memory PRM, and noise power N_(r0) stored in noise power estimator E(N_(r0)). [PRM and E(N_(r0)) are shown in FIG. 1.]. Matrix U is applied to an analyzer AYZ together with correlation vector y. Analyzer AYZ performs the function in Eq. (15) to obtain soft-output vector {tilde over (b)}.

Soft-output-vector {tilde over (b)} is applied to a hard decision circuit DEC to obtain a decision vector {circumflex over (b)}=[b₁, b₂, . . . b_(k), . . . b_(K)]^(T) consisting of detected values for K users. Above described function is performed by a basic circuit of a conventional detector, called MMSE-D, which is a partial circuit shown inside a dotted-line frame in FIG. 5.

Let us call the circuit above described the 0-th stage, MMSE-D by renaming the user-separating matrix as U⁰ and the soft-output vector as {tilde over (b)}⁰, and explain a function of a circuit shown outside the dotted-line frame.

CPE shown in FIG. 5 denotes a power estimator for matrix C defined by Eq. (17). Power estimator CPE calculates an evaluated power vector P_(C) ⁰=(P_(C1) ⁰, P_(C2) ⁰, . . . , P_(CK) ⁰)^(T) as the 0-th power for respective users, based on matrix U⁰. On the other hand, soft-output {tilde over (b)}⁰ is applied to a logarithmic likelihood ratio estimator LLRE to generate a normalized probability ratio λ⁰(=λ₁ ⁰, λ₂ ⁰, . . . λ_(K) ⁰) for the K users in Eq. (22). These output vectors P_(C) ⁰ and λ⁰ are applied to a best user decision circuit BUD. For example, a process expressed by Eq. (24-A) or (24-B) is performed to select one {tilde over (b)}_(k′) ⁰(k′ ε1, 2, . . . K) as the (first) best user soft-output among all the elements of soft-output vector {tilde over (b)}⁰. Selected soft-output {tilde over (b)}_(k′) ⁰ is applied to a hard decision device DEC to make a hard-decision output (detected value) {circumflex over (b)}_(k′) ⁰ (ε±1).

Detected value {circumflex over (b)}_(k′) ⁰ is applied to a modulator MOD so as to generate b_(k′) ⁰p_(k′). On the other hand, the best user k′ determined by best user decision circuit BUD is applied to a best user remover BUR to generate a removing vector −p_(k′). These outputs are applied to a summing circuit SU and user-separating matrix generator U-GEN to make a receive-vector r¹ _(c) as the first receive-vector and a pilot-response-matrix P¹ as the first pilot-response-matrix according to Eq. (25). These outputs r¹ _(C) and P¹, in addition to noise power N_(r0) are applied to the first stage MMSE-D to calculate a soft-output-vector {tilde over (b)}¹, and a user-separation-matrix U¹. Estimator LLRE produces a normalized probability ratio a normalized probability ratio λ¹ with {tilde over (b)}¹, and estimator CPE produces an evaluated power vector P¹ _(C). By using these results the best user decision circuit BUD determines {tilde over (b)}_(k′) ¹ (k″ ε1, 2, . . . K, k″≠k′) as the second best user soft-output, by performing the same operation as that done by the first stage MMSE-D. Then, decision circuit DEC obtains a decision value {circumflex over (b)}_(k′) ¹ for the k″-th user with soft-output {tilde over (b)}_(k′) ¹. Note that the number of candidate users for k″ is K−1, because the first receive vector r¹ _(C) dose not include the components of the first best user. These operations are repeated K times while using the k-th (k=0, 1, 2, . . . K−1) stage MMSE-D sequentially to generate decision outputs of all the users.

In this case, in place of a constant (noise power) N_(r0) of the additive vector N_(r0)I included in matrix U in Eq. (14), a generalized additive vector N_(r0)a given by the following equation can be used:

$\begin{matrix} \left. \begin{matrix} {U = {{P^{H}P} + {N_{r\; 0}a\; 1}}} \\ {a = \left\lbrack {a_{1},a_{2},\ldots \mspace{14mu},a_{K}} \right\rbrack} \\ {a_{k} > 0} \end{matrix} \right\} & (31) \end{matrix}$

According to the generalized additive vector, the value of the correct solution measure can be improved. Calculate correct solution measures obtained for a plurality of generalized additive vectors, then it is possible to find a user representing the best value among the correct solution measures, as the best user.

As an evaluation measure of system performance, let us use a power-bandwidth-product. This is defined as a product of a transmit-power P_(TX) and a transmission band-width B required for 1-bit transmission, as given by the following equation,

$\begin{matrix} \left. \begin{matrix} {\lbrack{PB}\rbrack = {\frac{P_{TX}B}{K} = \frac{\xi \; P_{BP}L\; \beta}{TK}}} \\ \left\lbrack {{P_{TX} = {\xi \; P_{BP}}},{\xi = {\left\lbrack {E_{b}/N_{0}} \right\rbrack_{SD}/\left\lbrack {E_{b}/N_{0}} \right\rbrack_{BP}}},{B = \frac{L\; \beta}{T}}} \right\rbrack \end{matrix} \right\} & (32) \end{matrix}$

where [E_(b)/N₀]_(BP) is a received SN ratio required to obtain error rate (for example, BER₀=10⁻³) when one user performs BPSK (Binary Phase Shift Keying) transmission under an AWGN (white noise) environment, P_(BP) is a transmit-power corresponding to [E_(b)/N₀]_(BP) (E_(b)/N₀ denotes a ratio of received energy E_(b) per bit and a received white noise power N₀ after despreading, and used as a theoretical reference value of the transmit-power), [E_(b)/N₀]_(SD) is a received SN ratio required to obtain the same value as BER₀ for the SD-MMSE system, β is a band-amplification-coefficient due to the guard sequences used in FIG. 3, and T is a core-symbol-period. Since the common terms can be set as P_(BP)=1 and T=1, Eq. (32) is simplified into the following equation:

$\begin{matrix} {\lbrack{PB}\rbrack = \frac{\xi \; L\; \beta}{K}} & (33) \end{matrix}$

Above-described sequential detection type MMSE (including DD) system, in comparison with the conventional system, can reduce considerably ξ, and K can be made equal to core-sequence-length L that is the maximum theoretically achievable value. For this reason, the value of [PB] can be considerably reduced, resulting in performance improvement.

An interference-control type MMSE detection system according to the second embodiment of the present invention will be described below.

In a minimum-mean-square-error detector MMSE-D, as expressed in Eq. (15), the soft-output includes interfering noise Δb₁. In order to keep the performance from this interfering noise, the interference-control type MMSE system is introduced.

An additive vector N_(r0)I in Eq. (14) can increase the regularity of matrix U and reduces the amplitude of white noise related error Δb_(x) depending on AWGN in Eq. (16-B). However, in compensation for the advantages, the interfering noise Δb_(I) in Eq. (16-C) generates. This is because components other than pilot-responses P constituting a receive-symbol is added to user separating matrix U due to the additive vector. In order to remove the interference generating function caused by the additive vector, it is effective to use Eq. (34-A) that is made by adding a term −N_(r0)b to the left-hand side of Eq. (13). Thus the interference removal can be achieved on the equation.

Ub−N _(r0) b=P ^(H) r _(C)  (34-A)

Ub=P ^(H) r _(C) +N _(r0) b  (34-B)

Making Eq. (34-B) by transposing term N_(r0)b to the right-hand side, and solving it with respect to b, then the following equation is obtained:

{tilde over (b)}=U ⁻¹ P ^(H) r _(C) +N _(r0) U ⁻¹ b  (35)

In the above equation, b in the right-hand side is an unknown, b cannot be used in the receiver. Therefore, let us consider a soft-output-vector of a detector MMSE-D as a soft-output of the 0th stage MMSE-D, denoted by {tilde over (b)}⁰. Using {tilde over (b)}⁰ as an approximate vector for b shown in the right-hand side of Eq. (35), a soft-output {tilde over (b)}¹ of the first stage MMSE-D can be generated as represented in the following equations,

{tilde over (b)} ¹ ={tilde over (b)} ⁰ +c ⁰  (36)

c ⁰ =U ⁻¹ N _(r0) {tilde over (b)} ⁰  (37)

Where c⁰ is a vector named as the 0-th interference-correcting vector.

Here, when it is assumed that core-receive-vector r_(c) does not include AWGN, a perfect correcting-vector is obtained when {tilde over (b)}⁰=b is satisfied in Eq. (37). For this reason, if the relation {tilde over (b)}¹=b is satisfied in Eq. (35), interfering noise can be completely removed. Considering a relation Δb≠0 indicated by Eq. (16), interfering noise cannot be completely removed. However, as long as {tilde over (b)}_(k) ⁰ b_(k)>0 is satisfied for the most of K components composing of vectors {tilde over (b)}⁰ and b, correcting vector c⁰ can display an effective correcting role. With this correction method, vector {tilde over (b)}¹ tends to be closer to transmit-vector b than {tilde over (b)}⁰.

When the first-stage correcting vector c¹ is calculated by using soft-output-vector {tilde over (b)}¹ of the first stage MMSE-D having a noise component less than that of the 0-th soft-output {tilde over (b)}⁰, a soft-output {tilde over (b)}² of the second stage MMSE-D using c¹ can be calculated from the following equations:

$\begin{matrix} \left. \begin{matrix} {{\overset{\sim}{b}}^{2} = {{\overset{\sim}{b}}^{0} + c^{1}}} \\ {c^{1} = {U^{- 1}N_{r\; 0}{\overset{\sim}{b}}^{1}}} \end{matrix} \right\} & (38) \end{matrix}$

The process is repeated n times to make a hard decision on a final soft-output {tilde over (b)}^(n) thereby obtaining a detected vector {circumflex over (b)}^(n), Where n can be set arbitrarily.

FIG. 6 is a diagram of the second embodiment of the present invention, showing a block diagram of an interference-control type minimum-mean-square-error detector. A minimum mean square error circuit (MMSE) shown in FIG. 6 is constituted by a partial circuit obtained by removing a portion of decision circuit DEC from detector (MMSE-D) inside a dotted-line frame in FIG. 5. A base-band receive-symbol r [r_(c)(t) in FIG. 3], a pilot-response-matrix P in Eq. (4), and a noise power N_(r0) contained in the receive-vector [an output of E(N_(r0)) in FIG. 1] are applied to circuit MMSE shown. Circuit MMSE generates a user-separating matrix U using matrix P and noise power N_(r0) and then produces a soft-output {tilde over (b)}⁰ of the 0-th stage MMSE and a user-separating matrix U according to the process of Eq. (15). These outputs are applied to an interference generator I-GEN₀. Generator I-GEN₀ generates interference-estimating vector c⁰ of the 0-th stage MMEE as the 0-th stage correcting term. By adding vector c⁰ to the 0-th soft-output {tilde over (b)}⁰, a soft-output {tilde over (b)}¹ of the first stage MMSE is obtained.

Soft-output-vector {tilde over (b)}¹ and matrix U which has been generated by the first stage MMSE are applied to an interference generator I-GEN₁ to generate an interference-correcting vector c¹ by the same method as that of generator IGN. The interference-correcting vector c¹ is added to the soft-output {tilde over (b)}₀ of the 0-th stage to generate {tilde over (b)}² of the second stage MMSE. By applying the n-th soft-output-vector {tilde over (b)}^(n) obtained after the processes of n stages to hard decision circuit DEC to make a final decision (detected) vector {circumflex over (b)}. In this manner, the transmit-data of the K users are decided. In Eq. (38), soft-output vector {tilde over (b)}¹ can also be used in place of vector {tilde over (b)}⁰ in the right-hand side.

The following means is used to further improve a correction-effect obtained by the correction vector.

(C-1): When the correction vector is used in an environment in which a received S/N ratio (E_(b)/N₀) is low, Eq. (14) is modified as:

U=P ^(H) P+λ _(N) N _(r0) I (λ_(N)>1)  (39),

As a coefficient λ_(N) is increased, Δb_(x) in Eq. (16-A) is reduced, and Δb_(I) is increased. The increased Δb_(I) is canceled by correction vector c.

(C-2): An amplitude of correction vector c is set as a function of a frequency (measure of interference intensity) such that respective components of vector {tilde over (b)} take values close to zero.

As this function, for example, the following equation is used as a threshold value a_(r)(ε0.5˜1).

$\begin{matrix} {{\theta \left( a_{r} \right)} = {\sqrt{\frac{P_{f}}{P_{b}}}\begin{bmatrix} {P_{f} = {E\left\{ \left( {1 - {{\overset{\sim}{b}}_{k}}} \right)^{2} \right\}}} & \left( {{{\overset{\sim}{b}}_{k}} < a_{r}} \right) \\ {P_{b} = {E\left( {{\overset{\sim}{b}}_{k}}^{2} \right)}} & \; \end{bmatrix}}} & (40) \end{matrix}$

where E denotes an operator taking an ensemble average.

(C-3): If a component |{tilde over (b)}_(k)|>1, in soft-output vector b takes a relation {tilde over (b)}_(k) when the correction vector c is used, excessive correction should have been performed. In order to avoid this problem, this system can introduce a threshold value B (<1), thereby modifies Eq. (37) so as to make the following equation:

$\begin{matrix} \left. \begin{matrix} {{c(B)} = {U^{- 1}\lambda_{N}N_{r\; 0}{\overset{\sim}{b}}^{L}}} & \; \\ {{\overset{\sim}{b}}_{k}^{L} = {\overset{\sim}{b}}_{k}} & {{{\overset{\sim}{b}}_{k}} \leq B} \\ {{\overset{\sim}{b}}_{k}^{L} = {{sign}\; \left( {\overset{\sim}{b}}_{k} \right)B}} & {{{\overset{\sim}{b}}_{k}} > B} \end{matrix} \right\} & (41) \end{matrix}$

When both the methods (C-2) and (C-3) are cooperatively used, the correction vector is given by the following equation:

c(a _(r) ,B)=θ(a _(r))U ⁻¹λ_(N) N _(r0) {tilde over (b)} ^(L)  (42)

The values λ_(N), a_(r), and B are selected so as to enhance the correction effect of interference-correcting vector c. This method can reduce a disturbing effect due to noise and interference which conventional MMSE method has to suffer, thereby can improve the error rate.

When the techniques of interference-correction according to the second embodiment is combined to the sequential detection type receiver of the first embodiment or third to fifth embodiments (will be described later), noise of these systems can be further reduced.

A combined pilot type decorrelating discrimination system according to the third embodiment of the present invention will be described below. In the first and second embodiments, an enveloped sequence obtained by adding a guard sequence to a core-sequence is used as the spreading-sequence. In this case, the receiver cannot use transmit-energy corresponding to the guard sequence length. As the data-rate increases, the guard sequence length to the core-sequence length ratio get too large to be neglected. In order to avoid this energy loss, the present invention, provides a system in which an user transmitter uses only a core-sequence to spread transmit-data, generating a transmit-symbol with the core-sequence, and a receiver demodulates a receive-symbol without suffering from ISI (inter-symbol interference).

FIG. 7 is an auxiliary explanatory diagram of the third embodiment, showing a time chart of base-band transmit- and receive-symbol-sequences with pilot-responses. Two upper rows in FIG. 7 show transmit-symbol-sequences s_(k)(n_(s)) from respective users u_(k)(k=1, 2) with a symbol-period T and a symbol number n_(s)=( . . . −1, 0, 1, 2, . . . ). Below these rows, a receive-data-symbol r is shown classified by multi-path-components. r_(k) is the k-th user specific receive-wave component corresponding to s_(k). In this example, the receive-wave component of each user consists of three waves, i.e., a direct wave, a T_(c) (chip period) delayed wave, and a 5T_(c) delayed wave.

The direct wave component of the n_(s)=(0)-th receive-symbol is represented by b_(k)(0)h_(k0)g_(k). Where g_(k) denotes a spreading-sequence of u_(k). With a delayed operator D, the components of the T_(c) and 5T_(c) delayed waves are expressed as b_(k)(0)h_(k1)g_(k)D and b_(k)(0)h_(k5)g_(k)D⁵, respectively. Here, b_(k)(n_(s)) denotes the (n_(s))-th data transmitted by user u_(k), and h_(kj)(J=0, 1, 2 . . . ) is the j-th component of a channel vector h_(k) from u_(k) to the receiver. The receiver extracts a sum of all the components (in this example, 6 waves) contained in the n_(s)-th receive-data-symbol r over a time position T_(s) (equal to symbol period T) and demodulates it in the symbol base. Period T_(s) includes the direct wave of a current symbol of interest and delayed-wave components of a preceding symbol.

It is assumed that only the transmitter of u₁ sends a pilot-symbol s_(P1) to the 0-th symbol position (period T, n_(s)=0) and does not send data or other pilot-symbols before and after the symbol position. Corresponding to s_(P1), as shown in the second bottom row, the receiver receives a pilot-response P₁ ^(R) which is a component enclosed by the real bulk frame. Since pilot-response P₁ ^(R) spreads over two-symbol-periods by the multi-path wave components. These components over symbol-timing zones of n_(s)=0 and n_(s)=1 are expressed as p_(1M) and P_(1D), respectively. (p_(1p) indicated by a dotted line denotes a delayed wave generated by an assumed preceding pilot, corresponding ti P_(1D).) Thus, response P₁ ^(R) is a receive-response when transmitting three transmit-data-symbols by setting as b₁(0)=p=1 and b₁(−1)=b₁(1)=0. (p denotes a pilot-information) Response P₂ ^(R) is a receive-response when a pilot is transmitted from the second user u₂. (In general, in order to prevent P₁ ^(R) and P₂ ^(R) from being simultaneously received, their transmit-timings are controlled.)

FIG. 8 is an explanatory diagram of the third embodiment of the present invention, showing a model diagram of pilot-responses.

In general, on an assumption that J multi-path-waves are received per transmit-symbol as a pilot-response, a receive-response is given by the following equation:

$\begin{matrix} {p_{k}^{R} = {\sum\limits_{j = 0}^{J - 1}\; {h_{kj}g_{k}D^{j}}}} & (43) \end{matrix}$

When a receive-symbol-synchronized position at the receiver is fixed to period T_(s), as shown in FIG. 8, response p^(R) _(k) consists of a main response p_(kM) on the current symbol-period and a delayed response P_(kD) on a subsequent symbol-period.

The components of receive-data-symbol r₁(0) over the n_(s)=0-th slot shown in FIG. 8 is given by:

r ₁(0)=b ₁(0)p _(1M) +b ₁(−1)p ₁ D  (44)

There are four combinations of two data b₁(0) and b₁(−1), if binary transmit-data is assumed. Let's use a combination of synthesized pilot-responses made by two element-responses for user u_(k) as shown in FIG. 8( a). This combination is given by the following equations.

p _(k) ⁺ =p _(kM) +p _(kD)=(p _(k1) ⁺ , p _(k2) ⁺ , . . . , p _(kL) ⁺)

p _(k) ⁻ =p _(kM) −p _(kD)=(p _(k1) ⁻ , p _(k2) ⁻ , . . . , p _(kL) ⁻)  (45)

Therefore, the (n_(s))-th receive-symbol r_(k)(n_(s)) received from user u_(k) can be expressed as follows:

r _(k)(n _(s))=b _(k) ⁺(n _(s))p _(k) ⁺ b _(k) ⁻(n _(s))p _(k) ⁻ (b_(k) ⁺,b_(k) ⁻ε+1,−1,0)  (46)

More specifically, when a combination, i.e., one pair of synthesized pilot-responses p⁺ _(k) and p⁻ _(k) are prepared, these responses include preceding symbol components. For this reason, a data-response can be analyzed without being subjected to an interference (ISI) due to a preceding symbol. For a synchronized position of the receive-symbol over period T_(s), the pilot-response received from user u₂ is delayed by τ₁₂ in case of FIG. 7. In this case, the pilot-response is generated as a sum of components as shown in FIG. 8( b).

In FIG. 8, an inter-user deviation τ_(kk′) (τ_(kk′) is an integer obtained by normalizing by T_(C)) of the pilot-responses (as indicated by τ₁₂) must satisfy the following equation if a synchronous or quasi-synchronous reception is assumed.

τ_(kk′)+(J−1)≦L (k≠k′)  (47)

Under the condition, the spreading range of pilot-response p^(R) _(k) is limited within a 2 symbol time-slot. [If p^(R) _(k) spreads in an n_(p) symbol-slots, a combination of synthesized pilot-responses of 2^(n) ^(p) ⁻¹ types should be prepared, instead of two types in the above example.]

In the example in FIG. 7, a system of user-separating equations for a receive-symbol r at position T_(s) can be expressed by the following equations, by using the same method as that in Eq. (11-A), with 2K synthesized pilot-responses.

$\begin{matrix} \begin{matrix} {r\; =} & {Pb} & + & x \\ \left\lbrack {L \times 1} \right\rbrack & {\left\lbrack {L \times 2\; K} \right\rbrack \left\lbrack {2\; K \times 1} \right\rbrack} & \; & \left\lbrack {L \times 1} \right\rbrack \end{matrix} & (48) \\ \left. \begin{matrix} {r = \left\lbrack {r_{1},r_{2},\ldots \mspace{14mu},r_{L}} \right\rbrack^{T}} \\ {P = \left\lbrack {p_{1}^{*},p_{2}^{*},{\ldots \mspace{14mu} p_{K}^{+}},p_{1}^{-},p_{2}^{-},{\ldots \mspace{14mu} p_{K}^{-}}} \right\rbrack} \\ {b = {\left( {b^{+} + b^{-}} \right) = \left\lbrack {b_{1}^{+},b_{2}^{+},{\ldots \mspace{14mu} b_{K}^{+}},b_{1}^{-},b_{2}^{-},{\ldots \mspace{14mu} b_{K}^{-}}} \right\rbrack^{T}}} \\ {x = \left\lbrack {x_{1},x_{2},\ldots \mspace{14mu},x_{L}} \right\rbrack^{T}} \end{matrix} \right\} & (49) \end{matrix}$

A soft-output-vector {tilde over (b)} is obtained by solving Eq. (48) with respect to b as in Eqs. (12) to (15).

The components of a data vector b(n_(s)) of the (n_(S))-th transmit-symbol are contained in not only a soft-output-vector {tilde over (b)}(n_(S)) that is a solution for the (n_(s))-th receive-symbol, but also a soft-output-vector {tilde over (b)}(n_(S)+1) of the subsequent receive-symbol. Therefore, concerned with the data components of user u_(k), the following relationship is satisfied:

{tilde over (b)} _(k)(n _(s))={tilde over (b)} _(k) ⁺(n _(s))+{tilde over (b)} _(k) ⁻(n _(s))+{tilde over (b)} _(k) ⁺(n _(s)+1)−{tilde over (b)} _(k) ⁻(n _(s)+1)  (50)

Accordingly, a data-corresponding soft-output can be calculated from a sum of two element soft-outputs obtained over two receive-symbols.

When a solution is calculated by a system of linear equations of a conventional DD or MMSE system, such as expressed by Eq. (15), the solution is given by {tilde over (b)}_(k)(n_(s))=b_(k)(n_(s))+Δb_(k)(n_(s)). Now, signal related elements which carry data b_(k)(n_(s)) in receive-vector r in FIG. 7 are four components, i.e., p_(kM)(n_(s)), p_(kD)(n_(s)), p_(kM)(n_(s)+1), and p_(kD)(n_(s)+1). For the simplicity, let's be an average voltage amplitude of vectors of these element components, T be a period of time, and E be energy. All the values are equal to each other. On the other hand, when projecting white noise components x(AWGN) included in r to main and delayed components p_(kM) and P_(kD) are represented by x/p_(kM) and x/p_(kD) respectively, a projected voltage component of x included in the right-hand side of Eq. (45) is given by [x/p_(kM)+x/p_(kD)]. Since the above-described two noises are not correlated to each other, all noise power is equal to a power-sum of the noises. When an average voltage amplitude of the noises is expressed by E|x/p_(kM)|=E|x/p_(kD)|=|x₀|, an SN ratio of a soft-output of Eq. (50) is given by the following equation:

$\begin{matrix} {{SN}_{k} = {\frac{\left( {4{s}} \right)^{2}}{2{x_{0}}^{2}} = {\frac{8{s}^{2}}{{x_{0}}^{2}} = \frac{8\; E}{N_{0}}}}} & (51) \end{matrix}$

When an enveloped sequence with the guard sequences in FIG. 3 is used, in the above explanation, p_(kD)=0 is established, and two symbol outputs cannot be used. For this reason, it should result an unfavorable relation SN_(k)=s²/x₀|². Accordingly, an increase in S/N ratio is achieved by using the combined pilot.

It is assumed that the multi-path propagation-characteristics shown in FIG. 7 is suddenly lost due to a change in the environment. In this case, the receiver always analyzes a known pilot-response, energies E_(kM) and E_(kD) of a main response and a delayed wave response are compared with each other. If the following equation holds good,

E_(kM)>>E_(kD)  (52),

the k-th pair of element vectors in matrix P in Eq. (48), is given by the following equation:

p _(k) ⁺ ≅p _(k) ⁻  (53)

For this reason, the regularity of the matrix P is considerably deteriorated. As a result, noise included in the soft-output-vector considerably increases.

For possible deterioration in the regularity of matrix P, an adaptive demodulation-function should be provided for a receiver. When the state described above is detected, the receiver forces to set, p_(kD)=0, p⁺ _(k)=p_(kM), and p⁻ _(k)=0 in Eq. (45) to reduce the size of matrix P in Eq. (49) to L×(2K−1), reducing the size of a vector b to (2K−1)×1 by removing b⁻ _(k) from vector b. And then it solves Eq. (48). Thus deterioration of regularity of matrix P can be avoided.

As the second means which does not especially require the change of the demodulating operation, an imitated delayed sequence system will be described below.

FIG. 9 is an auxiliary explanatory diagram of the third embodiment of the present invention, showing a diagram of transmit-symbols and pilot-responses of the imitated delayed wave transmission system. The n_(s)(=0, 1)-th transmit-symbols s_(k)(0) and s_(k)(1) of the k-th user are shown in upper rows in FIG. 9. At the tails of these symbols, additional symbols s⁰ _(k)(0) and S⁰ _(k)(1) are added as imitated symbols to the tail outside of a symbol-period T allocated in advance. For this reason, an extended spreading-sequence is given by:

ĝ _(k)=(g _(k1) , g _(k2) , . . . , g _(kL) , g _(k1) ^(o) , g _(k2) ^(o) , . . . , g _(kV) ^(o))  (54)

A sequence with V chips arranged tail outside the symbol-period is added to core-sequence g_(k) with L chips equivalent to the symbol-period as an imitated delayed sequence.

In the middle rows in FIG. 9, the same pilot-responses as in FIG. 7 are shown. p^(R) _(k)(h₁₀ to h₁₅) denotes a response constituted by six receive-waves, and p^(R) _(k)(h₁₀) denotes a response constituted by one receive-wave without delayed waves. (A waveform in FIG. 9 is modeled.)

The two bottom rows in FIG. 9 show a combination of synthesized pilot-responses expressed by Eq. (45) to cases in which there are no delayed wave.

Accordingly, even though there is no delayed wave, a relation p⁺ _(k)·p⁻ _(k) is satisfied, and the regularity of matrix P can be avoided from being deteriorated. For this reason, the demodulating process can be directly applied to the de-correlating system to obtain the solution.

Since the V chips of a header part of a symbol S_(k)(1) on a symbol period given to a slot n_(s)=1 in FIG. 9 has the same time zone as that of the V chips of a tail part of symbol s⁰ _(k)(0) that is a preceding symbol, both the chips overlap, and the transmitter has to make a sum of both the chips, and then transmit the sum. Let us consider of a combination of b_(k)(0) and b_(k)(1), such that the V chips of the both symbols may be partially or entirely canceled out by the summing. For this reason, in order to avoid a transmit-signal from being attenuated a sequence ĝ_(k) in Eq. (54) is desirably designed by a means which gives, for example, real amplitude to chips of gk₁ to g_(kv′) and imaginary amplitude to chips of g⁰ _(k1) to g⁰ _(kv) so that both the V chip sequences become orthogonal to each other.

The fourth embodiment of the present invention will be described below. The invention related to a multi-input multi-output (MIMO) CDMA system, provides a technique which achieves perfect inter-user interference separation to improve a diversity effect due to a number of input/output antennae.

FIG. 10 is a block diagram of a multi-input multi-output transceiver showing the fourth embodiment of the present invention. A transceiver TX_(k) of the k-th user u_(k) shown on the left in FIG. 10 transmits space-time signals Σ^(k) arranged in 4×4 dimensions through N transmit-antennae A^(T) _(n) (n=1, 2, . . . , N). A transmitter TX_(k′) of the similar k′(k′≠k)-th user (not shown) transmits a space-time signal Σ^(k)′.

A transmit-data-sequence [b^(k) _(d)] (d=1, 2, . . . N_(d) is an order number of data) of u_(k) is converted into N_(d)=4 parallel data by serial/parallel converter s-p. Outputs b^(k) _(d) of a data group constituted by the N_(d) parallel data modulate by a modulator MOD the same enveloped sequence e^(k) _(d) (including a core-sequence g^(k) _(d) as an element) as in FIG. 3 to generate a base-band transmit-symbol s^(k) _(d). Each symbol s^(k) _(d) is added to a space-time signal generator G_(d). In the space-time generator G_(d), a symbol-sequence S^(k) _(dτn) is generated by a delayed element D. Reference symbol τ(=1, 2, . . . , N_(τ)) denotes a time slot number, and reference numeral n(=1, 2, . . . , N) denotes a transmit-antenna number.

Accordingly, N_(d) space-time signals Σ^(k)(s^(k) _(τn)) are generated. Each symbol s^(k) _(dτn) is given by the following equation:

s_(dτn) ^(k)=b_(dτn) ^(k)g_(d) ^(k)  (55)

Here, a value b^(k) _(dτn) depends on only d. The symbol s^(k) _(dτn) which τ and n are equal to each other is synthesized with a transmit-symbol s^(k) _(τn)(t) by a summing gate SU. The transmit-symbol s^(k) _(τn)(t) [The transmit-symbol is assumed as an output generated by modulating a chip-waveform q(t) expressed in Eq. (1) and a carrier-wave] are sequentially transmitted from the n-th antenna A^(T) _(n).

The receiver RX will be described below. A signal r_(pnm)(t)[r_(mτ)(t)] received at a pilot [data] transmit-timing in a receive-symbol received through the m-th (m=1, 2, . . . , M, and M=2 in FIG. 10) receive-antenna A^(R) _(m) is added to a pilot-symbol demodulating circuit PD [data-symbol demodulating circuit SD]. Here, a base-band channel gain h^(k) _(τnm) [base-band data-symbol r_(mt)] is generated.

These base-band-signals are connected in series with each other through a delayed element D_(r) of one symbol-period (T_(E)) to obtain a synthesized pilot-response h^(k) _(τn) and a synthesized data-symbol r, N data-symbols r_(τ) are added to a receive-symbol generator RSG. In the receive-symbol generator RSG, a receive-symbol given by the following equation is generated.

$\begin{matrix} {r = {\sum\limits_{\tau = 1}^{N_{\tau}}\; {r_{\tau}D_{r}^{\tau - 1}}}} & (56) \end{matrix}$

where D_(r) is a delayed operator of one symbol.

On the other hand, a channel gain h^(k) _(τn) is added to a pilot-response generator PRG together with spreading-sequences g^(k) _(d) addressed to N_(d) used in each transmitter. In the pilot-response generator PRG a pilot-response-matrix P is generated. The pilot-response generator PRG estimates an AWGN included in a receive-symbol from a received pilot-signal r_(pnm)(t) to output a noise power N_(r0). The shown minimum-mean-square-error detector MMSE-D outputs a detected value {circumflex over (b)} of a data vector by the circuit in FIG. 5 by using the receive-symbol r, the pilot-response-matrix P, and the noise power N_(r0).

FIG. 11 is an auxiliary explanatory diagram of the fourth embodiment of the present invention and is an arrangement diagram of a space-time signal. This diagram shows the above-described space-time signal Σ^(k) in a two-dimensional space consisting of a time axis τ and a space axis n. The diagram shows a case in which N_(τ)=N=4. Each dotted-line frame determined by τ and n denote a transmit-timing and a transmit-antenna number allocated to the transmit-symbol s^(k) _(dτn). As reference symbols in the dotted-line frame in FIG. 11, k, n, and τ are omitted, and only s_(d) is shown. For this reason, a symbol which carries d-th data is allocated to each time slot and each antenna only once. A conventional space-time coding technique uses a system in which the symbol is multiplied by an element of an orthogonal code c, and c_(τn)s_(dτn) is allocated to the frame in FIG. 11. In this case, a single user is used as a target, each transmit-symbol is multiplied by c_(τn) and transmitted such that N_(d) transmit-symbols can be completely separated from each other by a receiver. At this time, on the receiving side, the d-th symbol is to be detected, and the receiver linearly sums N_(τ) receive-symbols to separate the d-th symbol from the N_(d) symbols.

Thereafter, a sum output is added to a matched filter which matches the sum output to the d-th pilot-response to obtain the soft-output. This technique can be operated on the assumption that a multi-user interference-separating function is not held and a channel-gain-characteristics is constant in a period of N_(τ) reception time slots, which is an abject of present invention. For this reason, there is a problem in which a diversity function achieved by interleaving (will be described later) cannot utilized.

A PD in FIG. 10 generates a channel gain h^(k) _(τnm) between transmit-/receive-antennae A^(T) _(n) and A^(R) _(m) which do not depend on a spreading-sequence used by the transmitter from the received pilot-signal r_(pnm)(t). When a channel gain h^(k) _(n) obtained by connecting h^(k) _(τnm) and a spreading-sequence g^(k) _(d) used by the k-th user in cascade in spreading of d-th data b^(k) _(d) are used, an analyzing pilot-response of the receive-symbol s^(k) _(dτn) is given by:

$\begin{matrix} {p_{d\; \tau \; n}^{k} = {\sum\limits_{m - 1}^{M}\; {p_{d\; \tau \; n\; m}^{k}D_{S}^{m - 1}}}} & (57) \\ {p_{d\; \tau \; n\; m}^{k} = {h_{\tau \; n\; m}^{k}\left\lbrack g_{d}^{k} \right\rbrack}^{T}} & (58) \end{matrix}$

When Eqs. (57) and (58) are used, the same de-correlating equation as in Eq. (11) is given by the following equation:

$\begin{matrix} {\left. \begin{matrix} {\overset{\;}{\overset{\begin{matrix} \; & \; & \mspace{20mu} \\ \overset{\Cap}{\tau} & \; & \; \end{matrix}}{\underset{\lbrack{N_{\tau}{ML} \times 1}\rbrack}{\begin{matrix} 1 \\ 2 \\ 3 \\ \underset{\Cup}{4} \end{matrix}\begin{pmatrix} r_{1} \\ r_{2} \\ r_{3} \\ r_{4} \end{pmatrix}}}\overset{\begin{matrix} \; & {r =} & \; & {Pb} & \; & + & \; & \times & \; \\ {(d} & 1 & {\; \;} & 2 & \mspace{11mu} & 3 & \mspace{31mu} & {4)} & \; \end{matrix}}{= \underset{{p^{1}{(u_{1})}}\mspace{14mu}\lbrack{N_{\tau}{ML} \times N_{d}K}\rbrack}{\begin{matrix} p_{111}^{1} & p_{212}^{1} & p_{313}^{1} & p_{414}^{1} \\ p_{122}^{1} & p_{221}^{1} & p_{324}^{1} & p_{423}^{1} \\ p_{133}^{1} & p_{234}^{1} & p_{331}^{1} & p_{432}^{1} \\ p_{144}^{1} & p_{243}^{1} & p_{342}^{1} & p_{441}^{1} \end{matrix}}}}\begin{matrix} \; \\ p^{2} \\ \left( u_{2} \right) \\ \; \end{matrix}\begin{matrix} \; \\ p^{3} \\ \left( u_{3} \right) \\ \; \end{matrix}\begin{matrix} \; \\ p^{4} \\ \left( u_{4} \right) \\ \; \end{matrix}\underset{\lbrack{N_{d}K \times 1}\rbrack}{\begin{pmatrix} b^{1} \\ b^{2} \\ b^{3} \\ b^{4} \end{pmatrix}}\underset{\lbrack{N_{\tau}{ML} \times 1}\rbrack}{\begin{pmatrix} x_{1} \\ x_{2} \\ x_{3} \\ x_{3} \end{pmatrix}}} \\ {{r_{\tau} = \left( {r_{r\; 1},r_{r\; 2}} \right)^{T}},{r_{\tau \; m} = \left( {r_{\tau \; m\; 1},r_{\tau \; m\; 2},\ldots \mspace{14mu},r_{\tau \; m\; L}} \right)^{T}},} \\ {{P^{k} = \left( {p_{1}^{k},p_{2}^{k},p_{3}^{k},p_{4}^{k}} \right)},{p_{d}^{k} = \left( {p_{d\; 1\; n}^{k},p_{d\; 2\; n^{\prime}}^{k},p_{d\; 3\; n^{\prime\prime}}^{k},p_{d\; 4\; n^{\prime\prime\prime}}^{k}} \right)^{T}}} \\ {p_{d\; \tau \; n}^{k} = \left( {p_{d\; \tau \; n\; 1}^{k},p_{d\; \tau \; n\; 2}^{k}} \right)^{T}} \\ {b^{k} = \left( {b_{1}^{k},b_{2}^{k},b_{3}^{k},b_{4}^{k}} \right)^{T}} \\ {x_{\tau} = \left( {x_{\tau \; 11},x_{\tau \; 12},\ldots \mspace{14mu},x_{\tau \; 1L},x_{\tau \; 21},x_{\tau \; 22},\ldots \mspace{14mu},x_{\tau \; 2L}} \right)^{T}} \end{matrix} \right\} \begin{bmatrix} {{d = 1},2,3,{4;{\tau = 1}},2,3,{4;}} \\ {{m = {1,2}},n,n^{\prime},n^{\prime\prime},{n^{\prime\prime\prime} \in {1 \sim 4}},{N_{d} = {N_{\tau} = {K = 4}}},{M = 2}} \end{bmatrix}} & (59) \end{matrix}$

Sizes are described under the respective matrices. In this case, in order to obtain a solution, K≦2L must be satisfied. In general, according to the number of dimensions [N_(τ)ML×N_(d)K] of P, the following expressions are obtained:

K≦N _(τ) ML/N _(d)  (60-A)

K≦ML (N_(τ)=N_(d))  (60-B)

As long as the above expressions are satisfied, Eq. (59) can be solved by a DD system. For this reason, complete symbol separation and user separation can be realized. When the equation is solved by using the principles of the first and second embodiments of the present invention as the MMSE system, a soft-output in which a sum of included interference and white noise is minimum is obtained. For this reason, an excellent error rate characteristic can be obtained. More specifically, according to the principles described above, highly-advance user separation can be realized.

In the example described above, an explanation is performed on the assumption that symbols s^(k) _(dτn) which carry the same data b^(k) _(d) to N_(τ) transmission time slots are arranged and transmission times of the time slots are adjacent to each other.

However, when the above equation is to be solved, unlike in the conventional technology, the channel-gain-characteristics need not be invariant [h^(k) _(τnm)=constant] with respect to τ. Therefore, a time diversity technique obtained by interleaving serving as effective means for performing radio transmission under a bad fading environment can be utilized. More specifically, in FIG. 11( a), four symbol sets transmitted by the r-th time slot are expressed by:

$\begin{matrix} {S_{i\; \tau} = {\left\{ {\sum\limits_{n = 1}^{4}\; s_{\tau \; n}} \right\}_{i} = \left( {s_{\tau \; 1},s_{\tau \; 2},s_{\tau \; 3},s_{\tau \; 4}} \right)_{i}}} & (61) \end{matrix}$

Reference symbol i denotes a combination number of a combination of N_(d) transmit-data. In the above example, N_(τ)=4 time slots are allocated to the i-th combination, and N=4 symbols are transmitted to each time slot. Eq. (61) is the τ-th transmit-symbol set of the i-th combination. (An uppercase k and a lowercase d are omitted in the equation.)

FIG. 11( b) is a write/read arrangement diagram for interleaving. In this case, transmit-symbol sets S_(iτ) are horizontally written in an order indicated by W like i=1, 2, 3, . . . Q, . . . . One S_(iτ) is constituted by N_(τ)=4 time slots (1, 2, 3, and 4). When a method of adding S_(Q+1, 1) next to the Q set and performing line feed is repeated five times, the example shown in FIG. 11( b) is obtained. When the symbol groups are sequentially read from the left in an order indicated by R, [S₁₁], S_(Q+1,2), S_(2Q+1,3), S_(3Q+1,4), and S_(4Q+2,1) are obtained. In this case, the column is changed to the right, the symbol groups continues as [S₁₂], S_(Q+1,3) . . . . This is an interleave output. Symbol sets separated from each other by 4Q symbol type are transmitted to adjacent time slots. On the receiving side, a reverse operation of the writing/reading operation (deinterleaving) is performed. At this time, receive-symbol set [R₁₁, R₁₂], . . . corresponding to [S₁₁, S₁₂], . . . are sequentially arranged. Therefore, the R₁₁ to R₁₄ are substituted into r₁, to r₄ of Eq. (59) to calculate a solution. By the interleaving, as channel-gain-characteristics h^(k) _(τnm) received by R₁₁ to R₁₄ in a transmission process, actual time interval between adjacent τ is 4QT. When Q is increased, correlation absence can be achieved. For this reason, anti-fading characteristic can be considerably improved.

When a coding technique using a known block code or a convolution-code is applied to the above system, the error rate can be further reduced. In this case, a transmitter performs coding to an input data sequence [b_(d) ^(K)] in FIG. 10 in advance to perform the above-described spreading modulation for converting the sequence into data sequence [b_(d) ^(KC)] and transmit the data sequence [b_(d) ^(KC)]. The receiver performs the above-described despreading demodulation to generate a decision output sequence [{circumflex over (b)}_(d) ^(KC)]. The receiver generates the decision output sequence [{circumflex over (b)}_(d) ^(KC)] by performing dispreading demodulation as described above and generates a detected value sequence [{circumflex over (b)}_(d) ^(K)] of original transmit-data by decoding the decision output sequence [{circumflex over (b)}_(d) ^(KC)]. However, in the above technique according to the present invention, the same data has been multiple-transmitted by N_(d)(=N_(Z)) symbols by using the space-time diversity, and an interleave function is added. For this reason, the technique includes a function of actually performing coding. Therefore, in comparison with a conventional system which does not have a sufficient user-separating function such as a single antenna transmission system using conventional coding and an interleaving function, a system having an considerably excellent power bandwidth characteristic can be realized.

The embodiment is based on the configuration of the MIMO system. In the embodiment, as described above, M receive-antennae are used to set a sequence length of transmit-symbols at L, so that a pilot vector length and receive-symbol-vector length can be set at ML.

Since this means that the number of dimensions of signals is M times, the maximum number K of users to be accommodated is advantageously increased by a factor of M, and noise is advantageously reduced to 1/M. This principle is applied to the first to fourth embodiments and the fifth embodiment (will be described later), so that a power-bandwidth-product PB serving as an evaluating measure for these systems further reduced advantageously.

A CDMA transmission system using an orthogonal sequence-set according to the fifth embodiment of the present invention will be described below.

As an example of an orthogonal sequence-set, a complementary sequence-set will be described below. When a sequence-set of two sequence combinations of A (A₁, A₂) and B(B₁, B₂), each consisting of a pair (in general, the number of powers of two) of the sequences with length L have a relationship expressed by the following four equations, this set is called a complete complementary sequence set.

$\begin{matrix} \left. \begin{matrix} {R_{AA} = {{{A_{1}*\overset{\_}{A_{1}}} + {A_{2}*\overset{\_}{A_{2}}}} = {\sum\limits_{n = 0}^{L - 1}\; {\rho_{A\; n}{\delta \left( {i - n} \right)}}}}} \\ {R_{BB} = {{{B_{1}*\overset{\_}{B_{1}}} + {B_{2}*\overset{\_}{B_{2}}}} = {\sum\limits_{n = 0}^{L - 1}\; {\rho_{B\; n}{\delta \left( {i - n} \right)}}}}} \\ {R_{BA} = {{{B_{1}*\overset{\_}{A_{1}}} + {B_{2}*\overset{\_}{A_{2}}}} = {\sum\limits_{n = 0}^{L - 1}\; {\rho_{C\; n}{\delta \left( {i - n} \right)}}}}} \\ {R_{AB} = {{{A_{1}*\overset{\_}{B_{1}}} + {A_{2}*\overset{\_}{B_{2}}}} = {\sum\limits_{n = 0}^{L - 1}\; {\rho_{D\; n}{\delta \left( {i - n} \right)}}}}} \end{matrix} \right\} & (62) \\ \left. \begin{matrix} {\rho_{A\; n} = {\rho_{B\; n} = 2}} & \left( {n = 0} \right) \\ {= 0} & \left( {n \neq 0} \right) \\ {\rho_{C\; n} = {\rho_{D\; n} = 2}} & \left( {n = 0} \right) \\ {= 0} & \left( {n \neq 0} \right) \end{matrix} \right\} & (63) \end{matrix}$

where *, -, and δ denote a periodical cross-correlation-function, a conjugate, and a delta function with a component variable i. n denotes a shift variable to indicate that the number of shifting a sequence is n. τ_(An) denotes the n-shift correlation, defined as a correlation between sequence A₁,(A₂) and the n-shift sequence of sequence Ā₁ (Ā₂), that is the n-th (n=0, 1 . . . L−1) component of correlation-function R_(AA). Therefore, it leads to R_(AA)=(ρ_(A0), ρ_(A1), . . . ρ_(AL−1)). As expressed in Eq.(63), let us use a sequence-set having both auto- and cross-orthogonal property except zero-shift auto-correlation. Although complete complementary sequence sets exist for a sequence length such as L=4, 8, 16, . . . , an example of a sequence-set consisting of sequences having a binary amplitude (+, −) and a length L=4 is described below:

$\begin{matrix} \left\{ \begin{matrix} \begin{matrix} {A_{1} =} & + & + & + & - \\ {A_{2} =} & + & - & + & +  \end{matrix} & \begin{Bmatrix} {B_{1} =} & + & + & - & + \\ {B_{2} =} & + & - & - & -  \end{Bmatrix} \end{matrix} \right. & (64) \end{matrix}$

Let us explain a composing method of transmit-symbols by using a combination of the orthogonal sequence pairs, and a receive-symbol components corresponding to the transmit-symbol.

FIG. 12 is an auxiliary explanatory diagram of the fifth embodiment of the present invention, showing a time chart of receive-/transmit-symbols using a complementary sequence-set in a CDMA system. The k-th user u_(k) (k=1, 2) transmits 2-bit data (b_(k) ^(A), b_(k) ^(B)) over a symbol period 2T_(E). Respective sequences A₁ ^(E) to B^(E) ₂ shown in the figure, denote enveloped (cyclic spreading) sequences with period T_(E) which are made by enclosing respective core-sequences over period T, that are sequences A₁ to B₂, with guard sequences of hatched portions shown in the figure. [Since the rear and front parts of each core-sequence are used as the front and rear guard sequence, a partial sequence with the consecutive L chips (corresponding to period T) of the enveloped sequence becomes a cyclic shift sequence of the core-sequence.] The enveloped sequence pair is multiplied by a data to produce a transmit-sub-symbol s₁ ^(A) that is represented as:

s ₁ ^(A) =b ₁ ^(A)(A ₁ ^(E) D ⊕A ₂ ^(E))  (65)

where ⊕ denotes a notation to arrange sequences A₁ ^(E) and A₂ ^(E) in cascade. A transmit-symbol of user u₁ is given by an arithmetic sum (addition in chip-element wise) of sub-symbols s₁ ^(A) and s₁ ^(B), where s₁ ^(B) is made by the same method as that of producing s₁ ^(A).

s ₁ =s ₁ ^(A) +s ₁ ^(B)=(b ₁ ^(A) A ₁ ^(E) +b ₁ ^(B) B ₁ ^(E))⊕(b ₁ ^(A) A ₂ ^(E) +b ₁ ^(B) B ₂ ^(B))  (66)

Although the components of A₁ ^(E) and B₁ ^(E) are summed up and then transmitted, these components can be separated and detected by a receiver due to the orthogonality between them. A transmit-symbol expressed by the following equation is generated for u₂ by the same method as described above.

s ₂ =s ₂ ^(A) +s ₂ ^(B)=(b ₂ ^(A) A ₁ ^(E) +b ₂ ^(B) B ₁ ^(E))⊕(b ₂ ^(A) A ₂ ^(E) +b ₂ ^(B) B ₂ ^(B))  (67)

where, D denotes a delay-operator by one chip. More specifically, symbol s₂ is composed of an enveloped sequence pair which is made using core-sequences obtained by shifting the core-sequences used for s₁ by one chip.

Let us consider synchronous reception in which these transmit-symbols arrive at the receiver at the same time. In FIG. 12, a user specific receive-symbol component corresponding to a transmit-symbol s_(k) is shown as r_(k)(r_(k) ^(A), r_(k) ^(B)). For convenience, FIG. 12 shows cases in which each receive-symbol consists of a direct wave component (h₁₀) and one delayed wave component (h₁₁) [The channel is assumed to be h₁=(h₁₀, h₁₁)]. Accordingly, a receive-symbol r₁ ^(A) has transmit-symbol (s₁ ^(A)) related components, and is given by the following equation, by assuming AWGN included in r₁ ^(A) to be 0:

r ₁ ^(A) =b ₁ ^(A)(h ₁₀ A ₁ ^(E) +h ₁₁ A ₁ ^(E) D)⊕b ₁ ^(A)(h ₁₀ A ₂ ^(E) +h ₁₁ A ₂ ^(E) D)  (68)

A delayed wave component of sequence A₁ ^(E) by n chips is represented as sequence A₁ ^(E)D^(n), that is obtained by multiplying the enveloped sequence by a delayed operator D^(n). A core-portion [r₁ ^(A)]_(C1) extracted from a front part of receive-symbol r₁ ^(A) is applied to a matched filter MF(A₁) matched to sequence A₁, and a core-portion [r₁ ^(A)]_(C2) extracted from a rear part of the receive-symbol is applied to another filter MF(A₂), and then a sum of both the correlated outputs is obtained. From the relationship of Eq. (63), the sum is represented as:

y ₁₀ ^(A) =[r ₁ ^(A)]_(C1) Ā ₁ +[r ₁ ^(A)]_(C2) Ā ₂ =b ₁ ^(A) h ₁₀  (69)

A receive-symbol r₁ ^(B) corresponding to a transmit-symbol s₁ ^(B) is given by the following equation based on Eq. (68):

r ₁ ^(B) =b ₁ ^(B)(h ₁₀ B ₁ ^(E) +h ₁₁ B ₁ ^(E) D)⊕b ₁ ^(B)(h ₁₀ B ₂ ^(E) +h ₁₁ B ₂ ^(E) D)  (70)

Since core-portion-sets [r₁ ^(B)]_(C1), [r₁ ^(B)]_(C2) and [r₁ ^(A)]_(C1), [r₁ ^(B)]_(C2) are orthogonal to each other according to Eq. (63), even though [r₁]_(C1), [r₁]_(C2) are used in place of [r₁ ^(A)]_(C1), [r₁ ^(A)]_(C2) in Eq. (69), an output y^(A) ₁₀ from the filter does not change. Therefore, the following equation holds good:

y ₁₀ ^(A) =[r ₁ ^(A)]_(C1) Ā ₁ +[r ₂ ^(A)]_(C2) Ā ₂ =b ₁ ^(A) h ₁₀  (71)

When two equal symbol components are applied to filters MF (A₁D) and MF(A₂D) matched to a delayed sequence, the following equation holds good:

y ₁₁ ^(A) =[r ₁ ^(A)]_(C1) A ₁ D +[r₁]_(C2) A ₂ D =b₁ ^(A) h ₁₁  (72)

If a receive-symbol r₂ is applied to matched filters MF (A₁D) to MF (B₂D²) by the same method as described above, the correlated outputs are represented as:

$\begin{matrix} \left. \begin{matrix} {y_{20}^{A} = {{{\left\lbrack r_{2} \right\rbrack_{C\; 1}\overset{\_}{A_{1}D}} + {\left\lbrack r_{2} \right\rbrack_{C\; 2}\overset{\_}{A_{2}D}}} = {b_{2}^{A}h_{20}}}} \\ {y_{21}^{A} = {{{\left\lbrack r_{2} \right\rbrack_{C\; 1}\overset{\_}{A_{1}D^{2}}} + {\left\lbrack r_{2} \right\rbrack_{C\; 2}{\overset{\_}{A_{2}D}}^{2}}} = {b_{2}^{A}h_{21}}}} \end{matrix} \right\} & (73) \end{matrix}$

In general, the j-th delayed wave related correlated output obtained with a receive-symbol from the k-th user u_(k) is given by the following equation:

$\begin{matrix} \left. \begin{matrix} {y_{kj}^{A} = {{{\left\lbrack r_{k} \right\rbrack_{C\; 1}{\overset{\_}{A_{1}D}}^{k - 1 + j}} + {\left\lbrack r_{k} \right\rbrack_{C\; 2}{\overset{\_}{A_{2}D}}^{k - 1 + j}}} = {b_{k}^{A}h_{kj}}}} \\ {y_{kj}^{B} = {{{\left\lbrack r_{k} \right\rbrack_{C\; 1}{\overset{\_}{B_{1}D}}^{k - 1 + j}} + {\left\lbrack r_{k} \right\rbrack_{C\; 2}{\overset{\_}{B_{2}D}}^{k - 1 + j}}} = {b_{k}^{B}h_{kj}}}} \end{matrix} \right\} & (74) \end{matrix}$

In general, the following equation holds good for a multiplexed receive-symbol including all the user-symbol-components,

$\begin{matrix} {r = {{\sum\limits_{k = 1}^{K}\; r_{k}} + x}} & (75) \end{matrix}$

When core-portions r_(c1) and r_(c2) (see FIG. 12) which are included in the front and rear parts of a multiplexed receive-symbol are applied to matched filters MF(A₁D¹) and MF(A₂D¹), respectively, correlated output vectors containing AWGN x are given by the following equations.

$\begin{matrix} {\left. \begin{matrix} {y_{l}^{A} = {{r_{C\; 1}\overset{\_}{A_{1}D^{l - 1}}} + {r_{C\; 2}\overset{\_}{A_{2}D^{l - 1}}} + x}} \\ {y_{l}^{B} = {{r_{C\; 1}\overset{\_}{B_{1}D^{l - 1}}} + {r_{C\; 2}\overset{\_}{B_{2}D^{l - 1}}} + x}} \end{matrix} \right\} \mspace{14mu} \left( {{l = {1,2}},\ldots \mspace{14mu},L} \right)} & (76) \end{matrix}$

As a consequence, the following equation holds good by referring to Eqs. (74) to (76).

$\begin{matrix} \left. \begin{matrix} {{H{\overset{\sim}{b}}^{A}} = y^{A}} \\ {{\begin{pmatrix} h_{10} & 0 & \ldots & h_{{K - 1},{J - 2}} & h_{K,{J - 1}} \\ h_{11} & h_{20} & \ldots & h_{{K - 1},{J - 1}} & 0 \\ h_{12} & h_{21} & \; & 0 & \vdots \\ \vdots & \vdots & \; & 0 & \vdots \\ h_{1,{J - 1}} & \vdots & \; & h_{{K - 1},0} & 0 \\ 0 & h_{2,{J - 1}} & \cdots & h_{{K - 1},l} & h_{K,0} \\ \vdots & 0 & \; & \vdots & h_{K,1} \\ \vdots & \vdots & \; & \vdots & \vdots \\ 0 & 0 & \; & h_{{K - 1},{J - 3}} & h_{K,{J - 2}} \end{pmatrix}\begin{pmatrix} {\overset{\sim}{b}}_{1}^{A} \\ {\overset{\sim}{b}}_{2}^{A} \\ \vdots \\ \vdots \\ \vdots \\ \vdots \\ \vdots \\ \vdots \\ {\overset{\sim}{b}}_{K}^{A} \end{pmatrix}} = \begin{pmatrix} y_{1}^{A} \\ y_{2}^{A} \\ \vdots \\ \vdots \\ \vdots \\ \vdots \\ \vdots \\ \vdots \\ y_{L}^{A} \end{pmatrix}} \end{matrix} \right\} & (77) \end{matrix}$

where, a channel gain vector is given by h_(k)=(h_(k0), h_(k1), . . . , h_(k,J−1)), and the number of users is given by K≦L. Similarly, for a receive-symbol component using a sequence-set B, the following equation holds good:

H{tilde over (b)}^(B)=y^(B)  (78)

These systems of linear equations can be solved with respect to {tilde over (b)}^(A) and {tilde over (b)}^(B) by using the first and second embodiments, so that data of K (≦L) users, each consisting of two bits, can be obtained. More specifically, when K=L is satisfied by using two core-sequences having 2L chips, the maximum data rate can be 2L bits per symbol. For this reason, high frequency-utilization-efficiency can be obtained.

In the above equation, since a diagonal component of matrix H corresponds to a dominant wave, as long as the amplitude of the dominant wave is large (when some dominant waves have a small amplitude, the transmit-timings of these users are controlled from a base-station, so that the dominant waves can shift to large-amplitude delayed waves), the regularity of matrix H can be kept high. Therefore, low error rate characteristics can be obtained without being easily disturbed by noise.

FIG. 13 is a block diagram of a receiver demodulating circuit according to the fifth embodiment of the present invention. The receive-symbols r_(c1) and r_(c2) shown in FIG. 12 are applied to matched filter banks MFB(A₁) and MFB(A₂), respectively.

Matched filters MF(A₁D^(l−1), and MF(A) ₂D^(l−1)) (l=1, 2, . . . , L) in these banks generate correlation outputs and generates an output in Eq. (76) through a subsequent summing circuit (SU). A soft-output-vector. y^(A)=(y^(A) ₁, y^(A) ₂, . . . , y^(A) _(L)), channel gain matrix H obtained by the pilot transmission in advance, and a noise power N_(r0) on the receive-symbol are applied to an analyzer AYZ. Analyzer AYZ generates a soft-output vector b ^(A) by using the circuit in FIG. 5 or 6.

The receive-symbol is also applied to matched filter banks MFB(B₁) and MFB(B₂) the same as the above matched filter banks. These outputs are applied to analyzer AYZ to generate a soft-output-vector b ^(B). A decision circuit DEC makes hard decisions on respective components of these soft-output vectors to obtain a detected data-vector {circumflex over (b)} with 2K-bit transmit-data.

As the maximum advantage of this system, Since a transmit-symbol is constituted of two sub-symbols made of sequence-sets A and B, the maximum peak transmit-power is four times larger than an element power of a system using only one symbol. On the other hand, in the conventional system, each user transmits a synthesized symbol which is made by summing up 2L sub-symbols composed of AD¹ and BD¹ (l=0, 1, 2, . . . , L−1), each uses one chip more delayed sequence than the sequence used by preceding one, to reduce a frequency band. This conventional system must be tolerate to the maximum peak transmit-power (2L)² times larger than the element power. However, the present invention provides a technique which avoids the above problem.

In the above explanation, synchronous reception is assumed. However, actually, even in quasi-synchronous reception, almost the same operation is performed.

This is because, if a spreading-sequence used for a user u_(k) is represented by AD^(n) and a position of a corresponding receive-symbol is delayed by 1 chip, a dominant wave and a j-chip delayed wave of the user specific receive-component is constituted by AD^(n+1) and AD^(n+j+1). As a result, the position of the k-th column vector h^(k) of matrix H in Eq. (77) is shifted to the next lower stage, and the diagonal component of matrix H includes 0. This degrades the regularity of matrix H. In this case, when a synchronized position of a receive-symbol is set at a position delayed by one chip, the matrix H has a configuration such that h^(k)′(k′≠k) is shifted to the next upper stage without changing h^(k). As a result, the problem of a diagonal component including 0 can be easily avoided.

In the above explanation, a complementary sequence-set is used as an orthogonal sequence-set. As another orthogonal sequence-set to be used for the present invention, there is a zero correction zone sequence-set (ZCZ). An example of a ZCZ sequence-set having a sequence length L=8 is described below:

$\begin{matrix} \left. \begin{matrix} \begin{matrix} {a_{0} =} & \left( + \right. & + & + & + & - & + & - & \left. + \right) \\ {a_{1} =} & \left( + \right. & + & - & - & - & + & + & \left. - \right) \end{matrix} \\ \begin{matrix} {a_{2} =} & \left( - \right. & + & - & + & + & + & + & \left. + \right) \\ {a_{3} =} & \left( - \right. & + & + & - & + & + & - & \left. - \right) \end{matrix} \end{matrix} \right\} & (79) \end{matrix}$

This example is constituted of four sequences for a family size M=4. A cyclic cross-correlation-function is given by:

$\begin{matrix} \left. \begin{matrix} {{\rho_{pq}(n)} = {\frac{1}{L}{\sum\limits_{i = 0}^{L - 1}\; {{a_{p}(i)}{a_{q}\left( {i - n} \right)}}}}} & \left( {p,{q \in {0,1,2,3}}} \right) \\ {\mspace{70mu} {= 1}} & \left( {{p = q},{n = 0}} \right) \\ {\mspace{70mu} {= 0}} & \left( {{p = q},{{n} \leq \Delta},{n \neq 0}} \right) \\ {\mspace{70mu} {= 0}} & \left( {{p \neq q},{{n} \leq \Delta}} \right) \end{matrix} \right\} & (80) \end{matrix}$

where Δ denotes a zero correlation zone. In the above example, Δ=1 is satisfied., Consider a set S_(z) constituted of L sequences (a₁ to a₄) belonging to a ZCZ sequence-set, and one-chip shifted sequence-set (a₁D to a₄D) and (a₁D⁻¹ to a₄D⁻²).

Any of cross-correlation between two arbitrary sequences in S_(z) takes 0. Therefore, each sequence of set S_(z) is allocated to each user, and the single sequence is used in place of a sequence pair obtained by shifting the complementary sequence pair (A₁, A₂) by an arbitrary number of shifts, so that the same system as the system using the complementary sequence-set can be constructed. Since a system of a K (=L) user can be designed by using a set having a sequence length L, the same frequency-utilization-efficiency as that the method using the complementary sequence-set can be achieved. For the same reason, a ZCZ sequence using complex or ternary values can be used for this purpose.

Conventionally, for user separation, a system using rows of an Hadamard matrix (Walsh function), so-called an orthogonal sequence, as spreading sequences has been practically used. For sequences p and q belonging to an Hadamard sequence-set, the n-shift auto-correlation takes a value of ρ_(pp)(n)≠0 (n≠0), and the n-shift cross-correlation takes a value of ρ_(pq)(n)≠0 (n≠0). More specifically, since shift correlation values of auto- and cross-correlation-functions take large values, this characteristic deteriorates the regularity of matrix H given by Eq. (77), resulting in degrading user-separating functions.

In contrast, the fifth embodiment of the present invention can completely remove the degradation, realize excellent user-separating characteristics, and reduce a power-bandwidth-product PB.

In the first to fifth embodiments, complete user separation or symbol separation can be performed by system MMSE-D described in the first to fifth embodiments. However, even though a DD system in which N_(r)=0 is set in Eq. (14), the same purpose can be accomplished. By designing the number K of users less than the core-sequence length (real spreading factor) L by a small percentage, system DD exhibits preferable characteristics with a satisfactory low error rate.

A high-performance pilot transmission system according to the sixth embodiment of the present invention will be described below. In order to cause a base-station receiver to sufficiently correctly recognize a channel-gain-characteristics between each user and a base-station, each user must transmit its pilot-symbol to the base-station at a frequency which is considerably higher than a Doppler frequency. Each user transmits the pilot in a time zone different from that of a pilot transmission and data transmission of other users, so that the receiver need to receive the pilot-symbol from each user without being subjected to interference and generate a sufficient correct pilot-response.

Furthermore, since one pilot-symbol generally includes large white noise (AWGN), a method which receives a plurality (N) of pilot-symbols within a short period of time T_(A) (time sufficiently shorter than a Doppler period T_(D)), integrates N obtained pilot-responses to reduce a power of the AWGN to a factor of 1/N. For this purpose, since KN pilot-symbol slots are set within a period of time T_(A), a time zone used in data transmission reduces. As a result, the reduction is a factor that deteriorates frequency-utilization-efficiency of the system. The embodiment to solve the problems described above will be described below.

FIG. 14 is an auxiliary explanatory diagram of the sixth embodiment of the present invention, and is a time chart of transmit- and receive-pilot-symbol-sequences.

In an upper section in FIG. 14, a user u_(k)(k=1, 2) transmits a transmit-pilot-symbol s_(pk) in a base-band band of the symbol-period T. A subsequent symbol slot is set in a blank slot as a guard section. (When a delayed wave of one or more symbols occurs, the blank slot is set to be long.)

Although the transmit-pilot-symbol s_(pk) has the same configuration as that of the data-symbol, carried information is given by p=1. It is assumed that the k-th spreading-sequence with a length of L chips is represented by g_(k) and that the k-th code-word (orthogonal sequence) of an orthogonal code having a length of an N chip is given by α_(k)=(α_(k1), α_(k2), . . . , α_(kN))^(T). In this case, the n-th transmit-pilot-symbol is given by:

s_(pk) ^(n)=pα_(kn)g_(k)=α_(kn)g_(k) (p=1)  (81)

(Data-symbol is generally b_(k)g_(k).)

In this case, it is assumed that a transmit-pilot-symbol set obtained by sequentially multiplying pilot-symbols (N pilot-symbols constitute one combination) by α_(kn) is expressed by:

S _(pk)=(s ¹ _(pk) , s ² _(pk), . . . , s^(N) _(pk))^(T)=(α_(k1) g _(k),α_(k2) g _(k), . . . α_(kN) g _(k))^(T)  (82)

It is assumed that, as an example of an orthogonal sequence α_(k), the k-th row of an Hadamard matrix having a size of N×N (N≧K).

The middle and lower sections in FIG. 14 show receive-symbols r_(pk) corresponding to s_(pk). The base-station controls transmit-timings of users such that receiving positions of both the symbols r_(pk) are equal to each other (synchronous reception) or fall within a range of a small timing difference (quasi-synchronous reception). When delay time of a delayed wave is roughly shorter than 1-symbol-period (T), a receive-pilot-symbol-period (T), a receive-pilot-symbol constituted by a main response α_(k1)p_(km) and a delayed wave response α_(k1)p_(kD) falls within a 2T symbol time-slot. Reference symbol r_(p2) in FIG. 14 shows a case in which a receive-timing is delayed by one chip. The pilot-symbols (N pilot-symbols constitute one set) are continuously transmitted (they can be spread between data-symbols), a receive-symbol-sequence of a period T_(p) is expressed as receiving sequences [r₁] and [r₂] in the lower section in FIG. 14. The period T_(p) is arranged in a middle of two adjacent data transmission periods (period T_(D)).

Since the receiver receives a synthesized wave of r₁ and r₂, a symbol of the n-th slot of a receive-pilot-symbol-sequence is generally given by the following equation:

$\begin{matrix} \left. \begin{matrix} {{r_{p}(n)} = {{\sum\limits_{k = 1}^{K}\; {\alpha_{kn}p_{k}}} + {x(n)}}} \\ {p_{k} = {p_{kM} + p_{kD}}} \end{matrix} \right\} & (83) \end{matrix}$

where x(n) is AWGN. It is supposed that a receive-response p_(k) is not variant in a period of the pilot period T_(p). At this time, when symbols r_(p) (n) of a receive-pilot-symbol set R_(p) are sequentially multiplied by the n-th element of α_(k) to generate a correlation sequence,

$\begin{matrix} {{\overset{\sim}{p}}_{k} = {{p_{k} + {\Delta \; p_{k}}} = {{\sum\limits_{n = 1}^{N}\; {\overset{\_}{\alpha_{k\; n}}{r_{p}(n)}}} + {x(n)}}}} & (84) \end{matrix}$

is obtained. This is a property of an orthogonal sequence. The property is a relationship calculated from the following equation:

$\begin{matrix} \left. \begin{matrix} {{\frac{1}{N}{\sum\limits_{n = 1}^{N}\; {\alpha_{k\; n}\overset{\_}{\alpha_{k^{\prime}n}}}}} = 1} & {k = k^{\prime}} \\ {= 0} & {k \neq k^{\prime}} \end{matrix} \right\} & (85) \end{matrix}$

In this case, the first term (signal component) of the right-hand side of Eq. (84) has a voltage sum, and the second term (AWGN) has characteristics of a power sum. For this reason, an S/N ratio of {tilde over (p)}_(k) is N times an S/N ratio obtained when a 1-pilot-symbol is received.

FIG. 15 is a block diagram of a transceiver of a orthogonal sequence modulation pilot transmission system according to the sixth embodiment of the present invention. A transceiver of the k-th user u_(k) prepares a sequence α_(k) of the k-th row of an Hadamard matrix. A transmitter TX_(k) uses the n-th element α_(kn) of the sequence α_(k) and causes a modulator MOD to modulate a spreading-sequence g_(k) having a blank symbol and to generate a transmit-pilot-symbol s_(pk)(n). N transmit-pilot-symbols s_(pk)(n) are sequentially generated and transmitted. (The carrier-wave modulating device is omitted in FIG. 15.)

A receiver RX(p_(k)) shows a diagram of a circuit unit which generates the k-th pilot-response. The receiver receives the n-th receive-pilot-symbol r_(p)(n) having a period of time of a period 2T in a pilot period. The modulator MOD multiplies the r_(p)(n) by the n-th component α_(kn) of an orthogonal sequence α_(k). The l-th element β_(k1) ^(n) of the multiplication output β_(k) ^(n)(=β_(k1) ^(n), β_(k2) ^(n), . . . , β_(k2L) ^(n))^(T) adds the l-th integrator Σ₁ in an integrating bank I_(k). After the operation is performed N times (n=1 to N), an integration value of each integrator is output as {tilde over (p)}_(kI). 2L outputs constitute a pilot-response-vector {tilde over (p)}_(k).

In this manner, a pilot period T_(P)=2T×N is shared by K users, time 2T×N'K required by a conventional system can be shortened to a factor of a value of 1/K.

An orthogonal sequence used in the above explanation may have a characteristic in which a 0-shift correlation value of two sequences in a set is 0. Therefore, not only a sequence based on the Hadamard matrix but also the complementary sequence, the ZCZ sequence, and the like can be used. In this manner, according to the present invention, the five embodiments are simultaneously used, frequency-utilization-efficiency can be further improved.

As described above, the invention described in claims 1 and 2 provides techniques of enhancing a user-separating function of a CDMA multi-user receiver using a minimum mean square error detector (MMSE-D) by introducing a correct solution measure directly related to the error powers contained in the respective soft-outputs which are obtained as solutions of a system of linear (de-correlating) equations with a user separating matrix, deciding a soft-output as the least error contaminated soft-output denoted by the best user, repeating the same method sequentially to modified systems of linear equations with reduced sizes to obtain the other user soft-outputs, and producing error reduced soft-outputs for all the users. As a result, it brings an effect of increasing the number of users to be accommodated in a CDMA system and reducing the required transmit-power, resulting in considerable reduction in a power-bandwidth-product PB of the system.

The invention described in claim 3 and 4 provides techniques of enhancing a user-separating function of a CDMA multi-user receiver using a minimum mean square error detector (MMSE-D) by introducing an interference-correcting term which is obtained with a soft-output vector of the 0-th stage (MMSE-D) as the 0-th soft-output vector {tilde over (b)}⁰ by assuming the vector to be of an errorless, producing another soft-out vector of the first stage MMSE-D as the first soft-output vector by adding the interference-correcting term to the 0-th soft-output vector, repeating a predetermined number of times the production of respective correcting terms and respective new soft-output vectors sequentially to produce a final soft-output vector with considerably reduced interference error components. Conventional MMSE systems with an interference compensating function have used an estimated interference disturbing one user of interest, which is obtained by estimated data-components of the other users, resulting in negligence of all the user components. For this reason, the improving effect of the conventional MMSE systems is very limited. The present invention brings an effect of increasing the number of users to be accommodated in a CDMA system and reducing the required transmit-power, resulting in considerable reduction in a power-bandwidth-product PB of the system.

The invention described in claims 5 and 12 provides techniques of reducing the required transmit-symbol-energy, by replacing each of the user specific pilot-response-vectors with a pair of synthesized pilot-response vectors which constitutes a pilot-response-matrix, establishing a system of linear equations with a user separating matrix composed of the pilot-response-matrix to avoid ISI (inter-symbol-interference), solving the system to produce a soft-output vector, deciding a current detected data-vector of a data symbol over a present symbol period, and generating transmit-data estimated by using a plurality of soft-output vectors of data symbols over present and adjacent symbol periods. Conventional multi-user receivers use guard sequence added envelope sequences to avoid ISI. However, as the data-rate increases, the delayed wave spread increases, resulting in that the guard sequence energy takes a dominant factor of symbol energy so as to lose the power efficiency. As a result, the present invention can achieve a required error rate with a low transmit-power even for a high data-rate transmission, resulting in considerable improvement in a power-bandwidth-product PB of the system.

The invention described in claims 6 and 7 provides techniques of increasing the number of users and reducing the error-rate by realizing a complete user-separating function and a data-separating function for a plurality of data symbols using plurality of space-time slots received at a receiver with an MIMO system, composing a user-separating matrix with user specific pilot response vector obtained via respective receive-antennae, and establishing a system of linear equations with the user separating matrix. In contrast to a frequency-utilization-efficiency of a conventional space-time coded transmission system with no user separation function, the present invention provides not only an increase in user population larger than the spreading factor, but also reducing the required transmit-power because of the diversity effect and an interleaving function. As a result, it brings an effect of increasing the number of users to be accommodated in a CDMA system and reducing the required transmit-power, resulting in considerable reduction in a power-bandwidth-product PB of the system.

The invention described in claim 8 provides technique to reduce considerably the transmit-power of a CDMA system by preparing a spreading sequence set whose member-sequence is made by shifting, by mutually different chips, a sequence belonging to a complementary sequence set or a zero correlation zone sequence set, allocating respective of the member-sequences to respective users as spreading-sequences, controlling the transmit-timing so that a receiver receives a receive-symbol under a synchronous or quasi-synchronous condition, analyzing the receive symbol with a system of linear equations with a user separation matrix having a high regularity due to the orthogonality of the member-sequences, and producing a soft-output vector without being disturbed by interfering noise and delayed waves. The techniques of claim 1 or 3 help reducing the error contained in the soft-outputs. As a result, in comparison with a conventional system in which each user producing a synthesized transmit-symbol made by adding multiple member-sequences, each conveying a data, present invention can considerably reduce the transmitter power while keeping almost the same frequency-utilization-efficiency.

The invention described in claim 9 provides a means for transmitting pilot symbols so that a receiver may obtain accurate pilot responses to identify the channels by allocating the k-th row of an Hadamard matrix as the k-th sequence to the k-th user, setting common pilot time slots where respective user-transmitters send out a pilot symbol sequence modulated by the k-th sequence, by using the same time zone and the same frequency band, so that the receiver effectively may utilize a time and a band occupied by all the pilot-symbol-sequences to separate perfectly respective user specific pilot responses. In order to have a correct pilot-response, conventional systems have utilized a transmission resource independently allocated for each user's pilot transmission, resulting in a considerable loss of the frequency-utilization-efficiency of the system. However, by adding the present invention to systems constructed based on the invention of claims 1 to 8, these systems can indicate an effect of further improving a power-bandwidth-product PB of the system.

The invention described in claim 10 provide a technique to simplify the systems equipped with an MMSE-D using a decorrelating detector (DD) in place of the MMSE-D described in claims 1 to 4. The present invention can indicate an excellent performance almost equal to that of the systems with MMSE-D by reducing the number of users to be accommodated by a small percentage.

The invention described in claim 11 provides techniques of increasing the number of users to be accommodated by introducing a concatenated receive-symbol-vector and a concatenated pilot-response-vector to a receiver of an MIMO system to increase the number of dimensions of a symbol to be analyzed, and to reduce the noise included in the soft-outputs. As a result, the present invention has an effect of considerably improve a power-bandwidth-product PB of the system.

The inventions described in claims 1 to 4 or claim 10 provides techniques of solving a general system of linear equations with multiple unknowns with an MMSE-D or a DD without being adversely affected by white noise and interference contained in an input vector. Therefore, these inventions can be applied to not only multi-user CDMA receivers described as examples, but also a tap coefficient control of channel equalizers. The inventions are widely applied for systems which need to solve a system of linear equations with multiple unknowns such as used for the field of communications, automatic control, and so on. 

1. A decorrelating discrimination system of code division multiple access signals, wherein a basic system structure is composed of plurality of cells, each of said plurality of cells comprises a base-station and K user-stations, each of the user-stations including a user transmitter and a user receiver, communicating through a multi-access-channel with the base-station which includes a base-station receiver and a base-station transmitter, and said user transmitter is capable of transmitting a data symbol to convey a data with a spreading sequence, and a pilot-symbol that is said spreading sequence to identify a channel from said user transmitter to said base-station receiver, and said base-station receiver includes a minimum mean square error detector to analyze an input vector, that is a receive-symbol containing both multiple user specific data responses, each having conveyed a transmit-data through a channel, in a way such that said minimum mean square error detector solves a system of linear equations with multiple unknowns made for said input vector, composed of a user separating matrix U consisting of a pilot matrix associated with said channels and a white noise power multiplied identity matrix, and an unknown data vector, characterized in that said receiver comprises: means for solving a system of equations as identified by the first system of decorrelating equations with a user separating matrix U⁰ to produce a soft-output vector {tilde over (b)}⁰ at an analyzing circuit, means for producing a variance of user-corresponding noise evaluation vectors of a noise evaluation matrix C generated by a matrix inverse of the user-separating matrix U as a correct solution measure P_(C) ⁰ consisting of K components at a power estimator, means for deciding one soft-output component {tilde over (b)}_(k′) ⁰ of said soft-output-vector {tilde over (b)}⁰ as the first best user u_(k′) based on one of the minimum candidate components of said correct solution measure P_(C) ⁰ at a best user decision circuit, means for making a hard decision on said soft-output {tilde over (b)}_(k′) ⁰ to obtain a detected value {circumflex over (b)}_(k′) ⁰ at a decision circuit, means for removing components corresponding to the first best user u_(k′) from the first system of decorrelating equations with said user separating matrix U⁰ with circuits of a modulator, a subtractor, and a best user remover to generate a system of equations as identified by the second system of decorrelating equations with a user separating matrix U¹, means for solving said second system to produce a soft-output vector {tilde over (b)}¹, producing a variance of user-corresponding noise evaluation vectors of a noise evaluation matrix C generated by a matrix inverse of said user separating matrix U¹ as a correct solution measure P_(C) ¹ consisting of (K−1) components, and deciding one output {tilde over (b)}_(k′) ¹ of the soft-outputs of said soft-output-vector {tilde over (b)}¹ as the second best user u_(k″) based on one of the minimum candidate components of said correct solution measure P_(C) ¹, means for making a hard decision on said soft-output {tilde over (b)}_(k′) ¹ to obtain a detected value {circumflex over (b)}_(k′) ¹, means for sequentially repeating the same method as that applied to the second system of decorrelating equations to the following systems of decorrelating equations, to decide the following best users, thereby producing said best users in turn, and means for making hard decision on soft-outputs of said best users to obtain detected values of transmit-data all the users have sent out at said decision circuit.
 2. A decorrelating discrimination system of code division multiple access signals, according to claim 1, characterized in that said receiver comprises: means for solving a system of equations as identified by the first system of decorrelating equations with a user separating matrix U⁰ to produce a soft-output vector {tilde over (b)}⁰ at an analyzing circuit, means for producing said variance of user-corresponding noise evaluation vectors P_(C) ⁰ consisting of K components at a power estimator, obtaining input noise power N_(r0) at a noise power estimator, and calculating a standard deviation σ⁰ to compose an error amplitude distribution with said variance P_(C) ⁰ and said input noise power N_(r0), means for obtaining a ratio identified by the 0-th normalized probability ratio λ⁰ consisting of K components that is calculated based on an error distribution model with said standard deviation σ⁰ and the K components of said soft-output vector {tilde over (b)}₀, means for deciding one soft-output component {tilde over (b)}_(k′) ⁰ of said soft-output-vector {tilde over (b)}⁰ as the first best user u_(k′) based on one of the maximum candidate components of normalized probability ratio λ⁰ at a best user decision circuit, means for making a hard decision on said soft-output {tilde over (b)}_(k′) ⁰ to obtain a detected value {circumflex over (b)}_(k′) ⁰ at a decision circuit, means for removing components corresponding to the first best user u_(k′) from the first system of decorrelating equations with said user separating matrix U⁰ according to claim 1, means for sequentially repeating the same method as that applied to the first system of decorrelating equations to the following systems of decorrelating equations, according to claim 1, and means for making hard decision on soft-outputs of said best users to obtain detected values of transmit-data all the users have sent out at said decision circuit.
 3. A decorrelating discrimination system of code division multiple access signals, wherein a basic system structure is composed of plurality of cells, each of said plurality of cells comprises a base-station and K user-stations, each of the user-stations including a user transmitter and a user receiver, communicating through a multi-access-channel with the base-station in the cell which includes a base-station receiver and a base-station transmitter, and a user transmitter is capable of transmitting a data symbol to convey a data with a spreading sequence, and a pilot-symbol that is said spreading sequence to identify a channel from said user transmitter to said base-station receiver, and said base-station receiver includes a minimum mean square error detector to analyze an input vector, that is a receive-symbol containing both multiple user specific data responses, each having conveyed a transmit-data through a channel, in a way such that said minimum mean square error detector solves a system of linear equations with multiple unknowns made for said input vector, composed of a user separating matrix U consisting of a pilot matrix associated with said channels and a white noise power multiplied identity matrix, and an unknown data vector, characterized in that said receiver comprises: means for solving a system of equations as identified by the first system of decorrelating equations with a user separating matrix U⁰ identified by the 0-th user separating matrix to produce a soft-output vector {tilde over (b)}⁰ identified by the 0-th soft-output vector at an analyzing circuit, means for multiplying the 0-th soft-output-vector {tilde over (b)}⁰ by a matrix inverse of the 0-th user-separating matrix U⁰ to calculate an interference-correcting vector c⁰ identified by the 0-th interference-correcting vector at an interference generator, and adding the 0-th interference-correcting vector c to the 0-th soft-output-vector {tilde over (b)}⁰ to produce a soft-output-vector {tilde over (b)}¹ identified by the first soft-output-vector, means for applying the same method to calculate an interference-correcting vector c¹ identified by the first interference-correcting vector using the first soft-output-vector {tilde over (b)}¹ as that used for calculating 0-th interference-correcting vector c⁰, means for applying and makes hard decisions on respective components of a soft-output-vector {tilde over (b)}_(n) of the n-th stage calculated by repeating a method of adding the first interference-correcting vector c¹ the 0-th soft-output-vector {tilde over (b)}⁰ to produce a soft-output vector {tilde over (b)}² identified by the second soft-output vector, means for n times repeating the same method as that applied to obtain the second soft-output vector {tilde over (b)}² to produce the n-th soft-output {tilde over (b)}^(n), and means for making hard decision on soft-outputs of the n-th soft-output {tilde over (b)}^(n) to obtain detected values of transmit-data all the users have sent out at said decision circuit.
 4. A decorrelating discrimination system of code division multiple access signals, according to claim 3, characterized in that said receiver comprises: means for introducing a coefficient λ_(N) to increase an amplitude of the identity matrix, used in the user-separating matrix U, and producing a system of decorrelating equations with a user-separating matrix U modified said coefficient λ_(N), means for limiting the amplitude of the soft-output-vector {tilde over (b)}⁰ of the 0-th stage calculated as a solution said system to produce a modified soft-output vector, generating an interference-correcting vector c⁰ of the 0-th stage by multiplying said modified soft-out vector by a matrix inverse of said user-separating matrix U, soft-out vector, means for adding a vector obtained by multiplying the interference correcting output c⁰ by an interference power estimated coefficient θ to the soft-output-vector {tilde over (b)}⁰ of the 0-th stage to generate a soft-output {tilde over (b)}¹ of the first stage, means for repeating the same method to the following stages to obtain a soft-output-vector {tilde over (b)}_(n) of the n-th stage, and, means for making hard decision on soft-outputs of the n-th soft-output {tilde over (b)}_(n) to obtain detected values of transmit-data all the users have sent out at said decision circuit.
 5. A decorrelating discrimination system of code division multiple access signals, according to claim 1 or 3, characterized in that said receiver comprises: means for receiving pilot-response-vectors received from respective user transmitters and separating each of them as a main response of a current pilot-symbol arrived on a target symbol-period and delayed wave responses of preceding pilot-symbols arrived on the same target symbol-period, and producing a pilot-response-set for each user, consisting of synthesized pilot-responses made by taking an algebraic sum of said main response and said delayed wave responses, means for generating a pilot-response-matrix P composed of said synthesized pilot-responses of all the users, generating a system of decorrelating equations with a user-separating matrix U made by said pilot-response-matrix P and an identity matrix, an unknown data-vector b, and receive-symbol-vector r as constituent elements, and solving said system according to a method of claim 1 or 3 to obtain a soft-output-vector.
 6. A decorrelating discrimination system of code division multiple access signals, according to claim 1 or 3, characterized in that, said basic system comprising: means for including a multiple-input multiple-output system in which a plurality of antennae are arranged to perform communications, each of said user transmitters comprising: means for allocating a plurality (N_(d)) of transmit-data to N_(τ)N symbols on a space-time transmit-axis constituted by a plurality (N_(τ)) time slots and a plurality (N) of transmit-antennae, and transmitting N_(τ)N symbols over N_(τ) symbol periods, and said base-station receiver comprising: means for receiving symbols over N_(τ) symbol slots at a plurality (M) of antennae, storing a pilot-response p_(dτnm) ^(k) of a pilot-symbol received at the m-th receive-antenna when the k-th user transmitter sends d-th transmit-pilot-symbol of N_(d) symbols over the τ-th symbol-slot of N_(τ) symbol-slots, generating a concatenated pilot-response-vector P_(d) ^(k) made by concatenating only pilot-responses p_(dτnm) ^(K) corresponding to the d-th pilot-responses with respect to antenna number m and time-sequence numbers τ, generating a pilot-response-matrix P consisting of these vectors, and generating a concatenated receive-vector r made by concatenating of M pieces of receive-symbol-vectors received on the N_(τ) symbol slots, means for generating a system of decorrelating equations with a user-separating matrix U generated from the pilot-response-matrix P and an identity matrix, the concatenated receive-vector r, and an unknown-data-vector b, means for solving said system of decorrelating equations according to claim 1 or 3 to obtain a soft-output vector {tilde over (b)} of the transmit-data-vector b, and making {tilde over (b)} hard decisions on respective components of the soft-output-vector {tilde over (b)} to obtain a detected data vector {circumflex over (b)}.
 7. A decorrelating discrimination system of code division multiple access signals, according to claim 6, characterized in that each of said user transmitters comprises: means for interleaving in advance a time sequence of N transmit-symbols where N is the number of transmit-antennae, and transmiting interleaved symbols over N_(τ) times, and said receiver comprises: means for performing deinterleaving M pieces of receive-symbols where M is equal to the number of receive-antennae, means for generating a system of decorrelating equations for each of N_(τ) symbol sets made by deinterleaved outputs, solving the system to obtain a soft-output-vector {tilde over (b)} of a transmit-data-vector b according to claim 6, and making hard decisions on respective elements of the soft-output-vector {tilde over (b)} to obtain a detected data-vector {circumflex over (b)}.
 8. A decorrelating discrimination system of code division multiple access signals according to claim 1 or 3, wherein a basic system structure is composed of plurality of cells, each of said plurality of cells comprises a base-station and K user-stations, each of the user-stations including a user transmitter and a user receiver, communicating through a multi-access-channel with the base-station which includes a base-station receiver and a base-station transmitter, and characterized in that said user transmitter comprises: means for transmitting a data symbol to convey a data with a spreading sequence, and a pilot-symbol that is said spreading sequence to identify a channel from said user transmitter to said base-station receiver, means for generating an enveloped cyclically shifted spreading-sequence made by adding guard sequences to a core-spreading-sequence which belongs to a k-shift sequence of one pair of complete complementary spreading-sequences or a k-shift sequence of a zero correlation zone spreading-sequence as said core-spreading-sequence, means for controlling the transmit-timing so that all of user specific receive-symbol components may arrive at said base-station receiver under a synchronous or quasi-synchronous condition, and said receiver comprises: means for extracting a core-period-part of the receive-symbol as an input vector, and analyzing it with a minimum mean square detector according to a method of claim 1 or
 3. 9. A decorrelating discrimination system of code division multiple access signals, according in any one of claims 1 to 4, characterized in that a user transmitter identified by the k-th user transmitter of K user transmitters comprises; means for generating a pilot-symbol with a guard added spreading-sequence, and preparing a pilot-symbol-sequence consisting of N symbols modulated by the k-th code-word with a code length N in an orthogonal code and transmitting said pilot-symbol-sequence so that it may arrive at the receiver together with other pilot-symbol-sequences sent out by the other user-stations under a synchronous or quasi-synchronous condition, and said base-station receiver comprises: means for receiving a pilot-response-sequence multiplexed by all of user specific pilot-responses, and applying said pilot-response-sequence to a matched filter matched to the k-th orthogonal code-word to generate a pilot-response-vector of the k-th user, and, means for producing a pilot-response-matrix P composed of pilot-response-vectors of all the K users to establish a said system of decorrelating equations used for respective claims 1 to
 8. 10. A decorrelating discrimination system of code division multiple access signals, according to any one of claims 1 to 4, characterized in that said receiver comprises: means for solving a system of equations as identified by the first system of decorrelating equations with a decorrelating detector which is made by removing an identity matrix I from a user separating matrix U used in said minimum mean square error detector and, means for solving said following systems of decorrelating equations with decorrelating detectors.
 11. A decorrelating discrimination system of code division multiple access signals, according claim 1 or 3, wherein said basic system comprising: means for including a multiple-input multiple-output system in which a plurality of antennae are arranged to perform communications, each of said user transmitters comprising: means for transmitting a data symbol to convey a data with a spreading sequence, and a pilot-symbol that is said spreading sequence to identify a channel from said user transmitter to said base-station receiver, and said base-station receiver comprising: means for receiving symbols over N_(τ) symbol slots at a plurality (M) of antennae, and characterized in that said receiver comprising: means for receiving M pieces of pilot-response-vectors per user obtained through M pieces of said antennae, generating an extended pilot-response-vector by concatenating said pilot-response-vectors and generating a pilot-response-matrix P by composing extended pilot-response-vectors obtained for all the users, means for generating an extended receive-vector r by concatenating all of the receive-symbols through M pieces of said antennae, establishing a system of decorrelating equations with a user separating matrix U made by said pilot-response-matrix P and solving said system to obtain a soft-output vector according to claim 1 or
 3. 12. A decorrelating discrimination system of code division multiple access signals, according to claim 5, characterized in that each of said user transmitters comprises: means for generating a data and a pilot-symbols with an extended sequence which is produced by adding an imitated delayed sequence to a core-spreading-sequence, so that the imitated delayed sequence is arranged outside the tail of a transmit-symbol-period that is the same time-slot as the core-spreading-sequence, and transmitting the data and the pilot-symbols so that a component corresponding to the imitated delayed sequence takes a time position overlapping a front portion of a subsequent symbol, transmitting a data-symbol and a pilot-symbol, and said receiver comprises: means for obtaining a receive-data-symbol and K user pilot-responses, and establishing a system of decorrelating equations with a user separating matrix or a pilot-response-matrix having an enhanced regularity, produced based on said receive data symbol and K user pilot-responses, and means for solving said system according to claim
 5. 